120 Design and analysis of differential passive circuits for I/Q generation in 60 GHz integrated circuits Ivan M. Milosavljević1,2, Dušan P. Krčum1,2, Lazar V. Saranovac1 1Department of Electronics, School of Electrical Engineering, University of Belgrade, Belgrade, Serbia 2NovelIC Microsystems, Belgrade, Serbia Abstract: I/Q generation circuit design is of great importance within the design of modern radars integrated on chip, as well as in high-speed communication circuits operating at millimeter wave frequencies. Due to increased robustness on process variations and noise, differential signaling and differential I/Q generators are of particular interest. Several passive topologies suitable for usage in the commercially available 130 nm SiGe BiCMOS process are presented and evaluated. These topologies are: branch-line coupler (BLC), broadside coupler (BSC), poly-phase filter (PPF) and quadrature all-pass filter (QAF). The first three topologies are implemented and the obtained results are compared to results published previously. The designed PPF has a total phase error of ± 2.5° over a 40 GHz bandwidth and this is the most desired solution if the total available area is a limiting factor. Transmission lines required for the design of BLC and BSC are small enough, making such structures easy to implement using today’s mainstream technologies. BLC is a reliable and widely used solution for I/Q generation at almost any microwave frequency. Designed BLC has a phase error of ± 1.7° over a 7 GHz bandwidth. BSC has proved to be the best solution for I/Q generation in the 60 GHz band. The designed solution has a smaller area than BLC, a phase error of only ± 5° over a 40 GHz bandwidth and ± 1° over a 7 GHz bandwidth. Keywords: millimeter-wave passive circuits; I/Q generation; 90o hybrid coupler; branch-line coupler; broadside coupler; poly-phase filter; quadrature all-pass filter Načrtovanje in analiza diferencialnih pasivnih vezij za generiranje I/Q v 60 GHz integriranih vezjih Izvleček: Načrtovanje I/Q generirnih vezij ime velik pomen v modernih integriranih radarjih, kakor tudi pri komunikacijskih vezjih visokih hitrostih v območji milimetrskih valovnih dolžin. Zaradi proizvodne robustnosti in šuma, so difencialni I/Q generatorji zelo pomembni. Predstavljene so številne pasivne topologije uporabne v komercialni 130 nm SiGe BiCMOS tehnologiji, in sicer: linijski sklopnik (BLC), hibridni sklopnik (BSC), polifazni filter (PPF) in kvadraturni polnoprepustni filter (QAF). Prve tri topologije so uporabljene in primerjane s prej objavljenimi rezultati. Načrtovan PPF ima skupno fazno napako ±2.5° pri 40 GHz pasovni širini in je najugodnejša rešitev, če imamo omejitve s prostorom. Prenosne linije, ki so potrebne za BLC in BSC so dovolj majne, da lahko ti topologiji uporabimo z uporabo današnjih tehnologij. BLC je zanesljiva in najbolj uporabljena rešitev v mikrovalovnih frekvencah. Fazna napaka načrtanega BLC je ±1.7° v 7 GHz pasovni širini. Najugodnejša rešitev v 60 GHz pasu je BSC. Potrebuje manjšo površino in ima fazno napako ±5° v 40 GHz pasu in ±1° v 7 GHz pasu. Ključne besede: milimeterska valovna pasivna vezja; I/Q generacija; 90° hibridni sklopnik; vejni sklopnik; polifazni filter; kvadraturni polnoprepusti filter * Corresponding Author’s e-mail: ivan.milosavljevic@novelic.com Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 120 – 129  MIDEM Society Review scientific paper 121 I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 1 Introduction Today’s millimeter-wave integrated circuits are pri- marily intended for communication and radar sensing applications. Advancements in silicon-based technol- ogy processes allowed the very high scale integration, making complete millimeter-wave systems integrated into a single chip. In these complex systems, usually there is a need for the generation of in-phase (I) and quadrature (Q) signals [1-5]. As an example, for blocks such as I/Q modulators [6], frequency multipliers [7], or vector-modulators [8-10], it is mandatory to have supporting circuits that provide the desired 90o phase shift between the input signals, in cases when quad- rature signals are not provided by the local oscillator (LO). Direct I/Q signal generation from the LO requires the design of a quadrature voltage controlled oscil- lator (QVCO). Although the design of a conventional parallel-coupled QVCO at millimeter-wave frequencies is feasible and reported in [11], the approach suffers from very high phase noise, and therefore its usage is limited. There are other QVCO state-of-the-art tech- niques based on injection- [12] and magnetic-coupling [13] that improve phase-noise, but their complexity and reliability issues rise with frequency. At microwave frequencies 90o phase generation can be provided by active quadrature dividers [14]. The use of dividers requires the VCO operating at twice the nominal fre- quency, therefore at higher millimeter-wave frequen- cies this is not a suitable topology. Another approach is using injection-locked frequency multipliers [15], that also suffer from similar problems as the approach with QVCO. Furthermore, using active circuits for I/Q gener- ation requires current-hungry and more sophisticated designs at millimeter-wave frequencies. In opposite to active circuits, passive circuits do not consume power. However, they introduce some losses that often need to be compensated, and the total loss per branch can- not be smaller than the theoretical minimum of 3 dB. Besides consumption, the key advantages of passive circuits are design simplicity and reliability. Conse- quently, a natural choice for I/Q signal generation at millimeter-wave frequencies is a passive circuit. There are various ways of generating quadrature sig- nals with passive circuits based on quarter-wave (λ/4) coupled and transmission line millimeter-wave struc- tures. The limiting factor for quarter-wave structures, also known in literature as 90o hybrid couplers, is chip area which is directly related to signal wavelength. This is the reason why these structures are much more in- teresting nowadays when the majority of high-speed communication systems operating frequencies are shifted to the millimeter-wave area. Passive quarter- wave structures for microwave frequencies have been investigated at various substrates. The investigations are mainly focused on miniaturization of quarter-wave coupled structures, as reported in [16, 17]. Most of them only deal with single-ended circuits, not suitable for integration in high-performance integrated circuits. Therefore this paper is focused only on topologies that have differential counterparts, in order to minimize sensitivity to noise and undesired couplings. On the other hand, passive circuits for quadrature generation are also designed from lumped components. The main representatives are RC and RLC based circuits and a vast number of papers comprised of circuits at millime- ter-wave frequencies have been reported [18-20]. The article is organized as follows: Section 2 shows the- oretical basics of different passive circuits for I/Q gener- ation. Implementation of circuits and simulation results obtained with electromagnetic (EM) solvers are shown in Section 3, followed by conclusions in Section 4. 2 Passive circuits for I/Q generation The main parameters of passive circuits for I/Q genera- tion are reflection coefficients, insertion loss, coupling, directivity, bandwidth, phase imbalance, amplitude imbalance, isolation between output ports and chip area. The priority of parameters is determined by spe- cific application. Typically in millimeter-wave integrat- ed circuits, the highest priority parameters are phase imbalance, insertion loss and chip area. Based on physical phenomena of quadrature genera- tion, passive circuits are divided in two main categories: I/Q generators based on distributed structures (trans- mission lines) and I/Q generators based on lumped components. 2.1 Distributed Transmission line-based passive circuits provide 90o phase shift using quarter-wave segments of transmis- sion lines. The most basic representative is the one with λ/4 transmission line inserted in the quadrature signal path. This approach requires a power divider for splitting the input signal in two signals of equal power. If a Wilkinson power divider [21] is used, sig- nificant chip area is required. In Figure 1, differential structures based on 90o transmission lines are shown. The input differential signal is LO, and the outputs are differential in-phase (I) and quadrature (Q) signals. Con- ventional approach shown in Figure 1 (a) suffers from narrowband operation, which can be improved using the Schiffman technique shown in Figure 1 (b) [22]. With this technique, performance is improved at the 122 expense of chip area. Both implementations are im- practical in millimeter-wave integrated circuits due to increased area and attenuation. Alternatively, distributed structures based on concen- trated coupling and coupled transmission lines are widely used in millimeter-wave integrated circuits. Figure 1: (a) 90o transmission line based power divider, (b) Schiffman 90o power divider. 2.1.1 Concentrated coupling Branch-line coupler (BLC) is a representative of trans- mission lines concentrated coupling. Conventional BLC hybrid is shown in Figure 2 (a). Since it occupies consid- erable chip area, reduction of transmission lines length is proposed in [16], by inserting shunt capacitance at the end of the transmission lines. This is shown in Fig- ure 2 (b). Figure 2: (a) Conventional branch-line coupler, (b) Re- duced size branch-line coupler. The relationships between electrical lengths of branch- line q1 and through-line q2, the differential character- istic impedance Z, the nominal differential impedance Zdiff, and shunt capacitance C in reduced-size BLC hy- brid are given as 1 , diffZarcsin Z θ   =    (1) 2 ,2 diffZarcsin Z θ   =    (2) 2 2 1 2diff diffdiff Z Z CZ Z Z ω     = − + −       . (3) In Figure 2 (b) it is given commonly used case when Zdiff /Z=1/√2, q1 = 30 o and q2 = 45 o. These values are very suited for practical implementations. Additionally, a technique of meandering is implemented whenever applicable to reduce total chip area. 2.1.2 Coupled line In microwave theory, I/Q generators based on coupled lines are known as directional couplers. Directional couplers are easily constructed from two λ/4 coupled transmission lines. The coupling effect is achieved by broadside coupling or edge-side coupling, as shown in Figure 3. The coupling ratio between two broadside coupled metal layers is given by 0 0 0 0 ,e o r air e o Z Zk k Z Z − = = × + ε (4) Where Z0e and Z0o are even- and odd-mode impedances of the coupled lines, er is the relative permittivity of the substrate, and kair is the coupling factor of the two lines in air. Figure 3: (a) Broadside coupling, (b) Edge-side cou- pling. In integrated realizations, the broadside coupler (BSC) is suitable for implementation, and its vertical ap- proach saves chip area. An important issue is that the distributed capacitance from the broadside coupled lines to the ground is asymmetric. Therefore, the width of transmission lines must be asymmetric to minimize asymmetry and optimize bandwidth, as shown in Fig- ure 3 (a). Alternatively, edge-side coupled structures shown in Figure 3 (b) can be used. They are very simple to de- sign, but suffer from several issues. Due to its inhomo- geneous dielectric nature, modal velocities of even- I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 123 and odd-modes are different. This difference leads to poor isolation and, consequently, worse directivity of the coupler. In order to overcome this problem, modal velocities should be equalized. Within practical solu- tions for equalization, the most suitable one for inte- grated circuit (IC) design is the placement of capacitors between input and output ports. However, even after applying this technique, overall performance of edge- side coupler remains mediocre. The edge-side coupler also suffers from reliability issues due to fabrication and tolerance problems. As a solution, multiple transmission lines are interdigi- tated in edge-side couplers. This structure is known as a Lange coupler [23], and the key design parameter is the voltage coupling coefficient c. Improved coupling helps in relaxing fabrication and tolerance problems. Millimeter-wave designs of single-ended unfolded Lange coupler are reported in [8, 24], etc. Differential Lange coupler requires two single-ended Lange struc- tures that significantly increase the area and complex- ity of the circuit, therefore they are rarely used. 2.2 Lumped components based Passive circuits for quadrature signal generation can be realized using basic circuit elements such as resistors, capacitors and inductors. Mentioned elements can eas- ily be implemented in currently available silicon-based processes. This approach in generation of quadrature signals leads to simple, inexpensive and very compact design. However, this design has large losses and a sig- nificant central frequency shift which is temperature and process dependent. Circuits for I/Q generation that use lumped elements are unable to handle larger pow- er levels. In practical designs, two main circuit topolo- gies are used: poly-phase filter (PPF) and quadrature all-pass filter (QAF). 2.2.1 Poly-phase filters PPF quadrature generation is based on phase shaping using combination of low-pass and high-pass RC filters. Low-pass (LPF) and high-pass filter (HPF) transfer func- tions are given by ( ) 1 , 1LPF H j j RC ω ω = + (5) ( ) . 1HPF j RCH j j RC ωω ω = + (6) Corresponding arguments are: ( ),LPF arctg RCθ ω= − (7) ( ). 2HPF arctg RCπθ ω= − (8) If the values of the components in LPF and HPF are cho- sen as 0 1RC ω = , where w0 is radial frequency of an input signal, phase shift through a LPF is - 45o and through HPF is + 45o, while attenuation is equal in both paths. The first order PPF achieves a narrowband quadrature generation and has a wide phase variation over process and temperature. More robust design can be obtained using higher order PPF. Second order PPFs are widely used, as shown in Figure 4. In order to reduce influence of the process variations, component values of each PPF stage are chosen for a different central frequency: 1 1 1 1 ,RC ω = (9) 2 2 2 1 .R C ω = (10) Relationship between the input radial frequency and poles of PPF is given by 2 1 2 0 ,ω ω ω= (11) where 1 0 2ω ω ω< < . Figure 4: Second order PPF schematic. 2.2.2 Quadrature all-pass filter QAF is used for phase shaping of quadrature output signals. In Figure 5, is shown the schematic of a fully differential QAF. Theory of operation is similar to PPF. Namely, QAF also uses combination of LPF and HPF for adequate phase shift between the output signals. Transfer functions be- tween LO input and I and Q outputs are given in equa- tions (12) and (13): I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 124 ( ) ,1I R j LH j R j L C ωω ω ω +=  + −   (12) ( ) 1 . 1Q R j CH j R j L C ωω ω ω + =  + −   (13) Figure 5: LC based QAF schematic. Phase difference between QAF outputs can be derived from the previous transfer functions, and after simplifi- cation the final expression for the QAF phase difference is: 2 1 .QAF LCarctg LRC R ωθ ω    +=     −     (14) In order to make the phase difference between I and Q signals equal to 90o, the resistance value should be ,LR C = (15) and for equal attenuation in both paths 2 0 1 .LC ω   =    (16) 3 Implementation and simulation results In this section, different passive I/Q generators are im- plemented using a commercially available 130 nm SiGe BiCMOS technology with seven metal layers shown in Figure 6. In all implementations solid metal 1 is used as circuit ground. All passive circuits are simulated in 2.5D planar EM solver based on method of moments (MoM), and verified in 3D full-wave solver based on finite element method (FEM). The comparison between simulation results is given for all designed circuits. The results ob- tained with MoM based simulator are shown as dashed lines, and the results obtained with FEM based simula- tor as solid lines. Impact of the mismatch and process variations of resistors and capacitors on phase and amplitude imbalance is simulated for all circuits using Monte Carlo simulations. Influence of the output load mismatch is also analyzed. Output impedances of both I and Q loads are varied in range 80% - 120% of nominal 100 Ω differential impedance. Effect of the mismatch and process variations and load mismatch is analyzed on single frequency of 60 GHz and characterized using ±3σ (standard deviation). 3.1 Branch-line coupler Implementation of differential reduced-size BLC is presented in this subsection. Transmission lines are implemented in 3 mm thick top metal 2. Differential impedance of branch-line and through-line is 141.4 Ω, which corresponds to lines of width 3 mm and spacing 17 mm. Transformation to a differential impedance of 100 Ω at input and output ports is achieved by reduc- ing line spacing to 7 mm and retaining same widths. According to equation (3), 43 fF differential capacitor value is obtained. Metal-insulator-metal (MIM) capaci- tors are used. In this technology, the bottom electrode of the MIM capacitor is connected to metal 5 and the Figure 6: Technology metal stack. Figure 7: 3D preview of the differential branch-line coupler. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 125 top electrode to top metal 1. Thus, two serial capacitors with twice the capacitance are used instead of a sin- gle differential capacitor. Capacitors are made in such a way to fit between differential lines and are connected in metal 5. 3D preview of the reduced-size BLC is shown in Figure 7. A meandering technique is used to reduce the overall BLC area, which is 267 mm x 230 mm. The termination port is connected to a 100 Ω resistor and the performance of the 60 GHz BLC obtained by MoM and FEM based simulators are shown in Figure 8. The BLC exhibits excellent performance in a 7 GHz bandwidth. Phase imbalance of the BLC is lower than 1.7o, and amplitude imbalance is lower than 0.4 dB. Reflection coefficients at input and outputs are lower than -14 dB. Performing process and mismatch Monte Carlo simulations gave us the insight in circuit’s be- havior under mentioned conditions. It is noticed that phase and amplitude imbalance does not vary more than 2.42º and 0.14 dB in the case of process and mis- match variations of the termination resistor and MIM capacitors. Major contributor on phase and amplitude imbalance deviation is the process variation of MIM capacitors and this should be taken into account in design process of reduced-size BLC. I/Q load mismatch effect on the phase imbalance is lower than 0.76º, while the amplitude imbalance is negligible. 3.2 Broadside coupler Differential input and output signals of the BSC are im- plemented in top metal 2 layer. Transmission lines are optimized to a differential impedance of 100 Ω. The width of lines is 3.5 mm and spacing is 4.5 mm. Coupled λ/4 lines are implemented in top metal 1 and metal 5 to achieve high coupling efficiency. Inter-layer distance is 0.85 mm. It requires the coupling coefficient k of 1/√2, and according to equation (4) values for even- and odd-mode impedances are approximately 241.5 Ω and 41.4 Ω. The widths of coupled lines are calculated from the EM simulation and top metal width of 5 mm and bottom metal width of 8 mm are obtained. In Figure 9, the geometry of 60 GHz differential BSC is shown. Ag- gressive meandering of the coupled lines is performed to minimize chip area. The BSC dimensions are 190 mm x 170 mm. Termination impedances T1 and T2 are chosen to be asymmetric in order to achieve very low phase mis- match. In this way the amplitude difference is con- sciously sacrificed. Termination impedance T1 is 20 Ω, and termination impedance T2 is 50 Ω. The perfor- mance of the 60 GHz BSC obtained by MoM and FEM simulators are shown in Figure 10. Figure 8: The performance of differential branch-line coupler: (a) phase difference, (b) insertion loss, (c) re- flection coefficient, (d) isolation. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 126 In the 40 GHz bandwidth, phase imbalance of the BSC is lower than 5o, and amplitude imbalance is lower than 2 dB. Reflection coefficients at inputs and outputs are lower than -10 dB, thus an excellent matching is achieved without any additional matching structures. Impact of the mismatch and process variations of ter- mination resistors on phase and amplitude imbalance is lower than 1.1o and 0.18 dB. Also, the load mismatch effect on the phase and amplitude imbalance is lower than 1.5o and 0.25 dB. 3.3 Poly-phase filter Two stage PPF is designed for 60 GHz band as fol- lows. Central operating frequency f1 for the first stage is 62.25 GHz, and for the second stage f2 is 59.25 GHz. Capacitance is chosen to be the same in both stages and equal to 39 fF. This is quite small value for the given technology, thus the series connection of four capaci- tors with capacitance of 156 fF is used. The resistance of the first stage is then R1 = 64.75Ω, and the resistance of the second stage is R2 = 68Ω. Dimensions of the designed PPF are 130 mm x 80 mm, and 3D preview is shown in Figure 11. The performance of the 60 GHz PPF are shown in Fig- ure 12. The phase imbalance is less than 2.5o at a 40 GHz range, but the PPF has significantly poorer matching. Impact of the mismatch and process variations of resis- tors and MIM capacitors on the phase and amplitude imbalance is lower than 0.6o and 0.1 dB making PPF very robust to mismatch and process variations. Load mismatch effect on the phase and amplitude imbal- ance is lower than 3.8o and 1 dB, which shows that the PPF is very sensitivity on output load mismatch. The performance summary is shown in Table 1. Figure 9: 3D preview of the differential broadside cou- pler. Figure 10: The performance of differential broadside coupler: (a) phase difference, (b) insertion loss, (c) re- flection coefficient, (d) isolation. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 127 Figure 11: 3D preview of the 60 GHz poly-phase filter. Figure 12: The performance of the 60 GHz poly-phase filter: (a) phase difference, (b) insertion loss, (c) reflec- tion coefficient, (d) isolation. 4 Conclusion An overview of I/Q generation circuits for the 60 GHz band is presented in this paper. The three most suit- able and most common topologies are chosen to be designed in a modern BiCMOS process. In comparison to other I/Q generation circuits, BSC ex- hibits the best performance. The main advantages of this design over other references is a small area and small phase error over a 40 GHz frequency range. In- sertion loss is comparable to theoretical values, thus it does not require an additional amplification stage. In a 7 GHz bandwidth around 60 GHz it presents supreme performance. All of these parameters highlight BSC as the first choice for an I/Q generation block in modern CMOS designs, in millimeter-wave integrated circuits. A possible drawback of this design is the usage of a specific metal stack. Depending on the vertical spacing between metal layers used for the implementation of the coupler, desired even- and odd-mode impedances may not be achievable. On the other hand, BLC is implemented in the same metal layer and does not have the technology related drawbacks. Small areas can be obtained using mean- dering and capacitive termination of λ/4 lines. How- ever, using capacitive termination of λ/4 lines leads to increased phase and amplitude deviation related with process and mismatch variations of used capacitors. Phase error in a 7 GHz bandwidth around 60 GHz is no- ticeably low and all ports are well matched. Insertion loss and output isolation are equally good as for BSC. However, BLC occupies a larger area than BSC. A poly-phase filter is used as the third I/Q generation circuit for applications in the 60 GHz band. This solution can be used in situations where area is critical. 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Reference Type Pol.a BW [GHz] |Sii|b [dB] |Sij|c [dB] |S32| [dB] PI [0] AI [dB] Area [mm2] [1] BLC S.e. 7 -11 -6 -14 ± 3.6 ± 0.75 370x270 [24] Lange S.e. 60 -16 -4.5 -14 ± 9.6 ± 2.25 300x160 [24] Lange S.e. 60 -13 -4.5 -11 ± 11.5 ± 1.75 160x120 [5] BSC S.e. 30 -15 -4.5 N/A ± 3 ± 0.75 280x200 [6] BSC Diff. 40 -13 -5 N/A ± 5.3 ±0.6 350x175 [25] BLC Diff. 7 N/A N/A N/A ± 4 ± 0.5 400x550 [18] PPF Diff. 2 N/A -10 N/A ± 2.1 N/A 210x200 This workd BLC Diff. 7 -14 -4.5 -17 ± 1.7 ± 0.4 267x230 This workd BSC Diff. 40 -10 -4.5 -15 ± 5 ± 1.5 190x170 This workd PPF Diff. 40 -5 -14 -22 ± 2.5 ± 0.75 130x80 a polarity (Pol.): single ended (S.e.) or differential (Diff.) b input and output reflection coefficients: worst or any known values c insertion loss: average or any known value d simulation results I. M. 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