Review scientific paper Original scientific paper Original scientific paper 121 130 113 © MIDEM Society 115 148 111 149 114 © MIDEM Society © MIDEM Society 120 112 ISSN 0352-9045 Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), September 2016 Revija za mikroelektroniko, elektronske sestavne dele in materiale letnik 46, številka 3(2016), September 2016 UDK 621.3:(53+54+621+66)(05)(497.1)=00 ISSN 0352-9045 Informacije MIDEM 3-2016 Journal of Microelectronics, Electronic Components and Materials VOLUME 46, NO. 3(159), LJUBLJANA, SEPTEMBER 2016 | LETNIK 46, NO. 3(159), LJUBLJANA, SEPTEMBER 2016 Published quarterly (March, June, September, December) by Society for Microelectronics, Electronic Components and Materials - MIDEM. Copyright © 2016. All rights reserved. | Revija izhaja trimesečno (marec, junij, september, december). Izdaja Strokovno društvo za mikroelektroniko, elektronske sestavne dele in materiale – Društvo MIDEM. Copyright © 2016. 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Editor in Chief | Glavni in odgovorni urednik Marko Topič, University of Ljubljana (UL), Faculty of Electrical Engineering, Slovenia Editor of Electronic Edition | Urednik elektronske izdaje Kristijan Brecl, UL, Faculty of Electrical Engineering, Slovenia Associate Editors | Odgovorni področni uredniki Vanja Ambrožič, UL, Faculty of Electrical Engineering, Slovenia Arpad Bürmen, UL, Faculty of Electrical Engineering, Slovenia Danjela Kuščer Hrovatin, Jožef Stefan Institute, Slovenia Matija Pirc, UL, Faculty of Electrical Engineering, Slovenia Matjaž Vidmar, UL, Faculty of Electrical Engineering, Slovenia Editorial Board | Uredniški odbor Mohamed Akil, ESIEE PARIS, France Giuseppe Buja, University of Padova, Italy Gian-Franco Dalla Betta, University of Trento, Italy Martyn Fice, University College London, United Kingdom Ciprian Iliescu, Institute of Bioengineering and Nanotechnology, A*STAR, Singapore Malgorzata Jakubowska, Warsaw University of Technology, Poland Marc Lethiecq, University of Tours, France Teresa Orlowska-Kowalska, Wroclaw University of Technology, Poland Luca Palmieri, University of Padova, Italy International Advisory Board | Časopisni svet Janez Trontelj, UL, Faculty of Electrical Engineering, Slovenia - Chairman Cor Claeys, IMEC, Leuven, Belgium Denis Đonlagić, University of Maribor, Faculty of Elec. Eng. and Computer Science, Slovenia Zvonko Fazarinc, CIS, Stanford University, Stanford, USA Leszek J. Golonka, Technical University Wroclaw, Wroclaw, Poland Jean-Marie Haussonne, EIC-LUSAC, Octeville, France Barbara Malič, Jožef Stefan Institute, Slovenia Miran Mozetič, Jožef Stefan Institute, Slovenia Stane Pejovnik, UL, Faculty of Chemistry and Chemical Technology, Slovenia Giorgio Pignatel, University of Perugia, Italy Giovanni Soncini, University of Trento, Trento, Italy Iztok Šorli, MIKROIKS d.o.o., Ljubljana, Slovenia Hong Wang, Xi´an Jiaotong University, China Headquarters | Naslov uredništva Uredništvo Informacije MIDEM MIDEM pri MIKROIKS Stegne 11, 1521 Ljubljana, Slovenia T. +386 (0)1 513 37 68 F. + 386 (0)1 513 37 71 E. info@midem-drustvo.si www.midem-drustvo.si Annual subscription rate is 160 EUR, separate issue is 40 EUR. MIDEM members and Society sponsors receive current issues for free. Scientific Council for Technical Sciences of Slovenian Research Agency has recognized Informacije MIDEM as scientific Journal for microelectronics, electronic components and materials. Publishing of the Journal is cofi­nanced by Slovenian Research Agency and by Society sponsors. Scientific and professional papers published in the journal are indexed and abstracted in COBISS and INSPEC databases. The Journal is indexed by ISI® for Sci Search®, Research Alert® and Material Science Citation Index™. | Letna naročnina je 160 EUR, cena posamezne številke pa 40 EUR. Člani in sponzorji MIDEM prejemajo posamezne številke brezplačno. Znanstveni svet za tehnične vede je podal pozitivno mnenje o reviji kot znanstveno-strokovni reviji za mikroelektroniko, elektronske sestavne dele in materiale. Izdajo revije sofinancirajo ARRS in sponzorji društva. Znanstveno-strokovne prispevke objavljene v Informacijah MIDEM zajemamo v podatkovne baze COBISS in INSPEC. Prispevke iz revije zajema ISI® v naslednje svoje produkte: Sci Search®, Research Alert® in Materials Science Citation Index™. Design | Oblikovanje: Snežana Madić Lešnik; Printed by | tisk: Biro M, Ljubljana; Circulation | Naklada: 1000 issues | izvodov; Slovenia Taxe Percue | Poštnina plačana pri pošti 1102 Ljubljana Journal of Microelectronics, Electronic Components and Materials vol. 46, No. 3(2016) Content | Vsebina 113 120 130 136 142 149 154 Pregledni znanstveni članek S. Pevec, B. Lenardič, D. Đonlagić: Mikroobdelava povsem vlakenskih fotonskih mikrostruktur za področje mikrofluidnih aplikacij I. M. Milosavljević, D. P. Krčum, L. V. Saranovac: Načrtovanje in analiza diferencialnih pasivnih vezij za generiranje I/Q v 60 GHz integriranih vezjih Izvirni znanstveni članki W. Tangsrirat: Dvojni sinusni kvadrantni oscillator z enosnim CC­CTA in ozemljenimi kondenzatorji T. Pečnik, S. Glinšek, B. Kmet, B. Malič: Tankoplastni kondenzatorji na osnovi Ba0.5Sr0.5TiO3, pripravljenega s sintezo v raztopini, s konfiguracijo kovina - dielektrik - kovina R. Kimovec, H. Helmers, A. W. Bett, M. Topič: Vpliv temperature in vzbujalnega toka na elektro­luminiscenco enosegmentnih enospojnih pretvornikov laserske moči M. Podhraški, J. Trontelj: Sistem z mikrotransformatorji za inkrementalno merjenje linearnega pomika M. Makarovič, J. Walker, E. Khomyakova, A. Benčan, B. Malič, T. Rojac: Kontrola električne prevodnosti v feroelektrični keramiki 0.7BiFeO3-0.3SrTiO3 z žganjem v dušikovi atmosferi in dopiranjem z Mn Naslovnica: Slika prostorske elektroluminiscence šest-segmentnega pretvornika laserske moči narejenega iz GaAs (Kimovec et al.) Review scientific paper S. Pevec, B. Lenardič, D. Đonlagić: Micromachining of All-Fiber Photonic Micro-Structures for Microfluidic Applications I. M. Milosavljević, D. P. Krčum, L. V. Saranovac: Design and Analysis of Differential Passive Circuits for I/Q Generation in 60 GHz Integrated Circuits Original scientific paper W. Tangsrirat: Dual-Mode Sinusoidal Quadrature Oscillator with Single CCCTA and Grounded Capacitors T. Pečnik, S. Glinšek, B. Kmet, B. Malič: Solution-derived Ba0.5Sr0.5TiO3 Thin-film Capacitors in Metal-insulator-metal Configuration R. Kimovec, H. Helmers, A. W. Bett, M. Topič: Temperature and Injection Current dependent Electroluminescence for Evaluation of Single-Junction Single-Segment GaAs Laser Power Converter M. Podhraški, J. Trontelj: Linear Incremental Displacement Measurement System with Microtransformers M. Makarovič, J. Walker, E. Khomyakova, A. Benčan, B. Malič, T. Rojac: Control of Electrical Conductivity in 0.7BiFeO3-0.3SrTiO3 Ferroelectric Ceramics Via Thermal Treat­ment in Nitrogen Atmosphere and Mn Doping Front page: Spatial electroluminescence image of six-segment GaAs laser power converter (Kimovec et al.) Editorial | Uvodnik Dear Reader, In summer all journals listed in SCIE eagerly waited for the Journal Citation Reports® (JCR) by Thompson Reuters and journals’ impact factor (IF) values for 2015. In early autumn also data for SNIP (Source Normalized Impact per Paper) — one of several metrics available on ScienceDirect — in 2015 came out. IFs are counted as a metrics that provide a snapshot of performance and help establish benchmarks for growth. We are pleased to see that metrics for our journal is improving: JCR IF (2015) = 0,433 and ScienceDirect SNIP (2015) = 0,393. Editorial Board is pleased that efforts and dedication to high quality harvest success. Associate Editor Professor Slavko Amon decided to step down due to retirement. We are truly grateful for his 14 years of dedicated work and contribution to the success of the journal. Associate Editor Professor Andrej Žemva decided to step down due to too many duties, especially since he is the Editor-in-Chief of another scientific journal Elektrotehniški vestnik. We are truly grateful for his 5 years of dedicated work and contribution to the success of the journal. A new member of Editorial Board, Professor Arpad Buermen has been elected by Executive Board of the MIDEM Society in June 2016. He will serve as the Associate Editor for Electronics. Please join the members of Editorial Board of the Informacije MIDEM - Journal of Microelectronics, Electronic Compo­nents and Materials in thanking Prof. Slavko Amon and Prof. Andrej Žemva for their valuable contribution and congratu­lating Prof. Arpad Buermen on his new position as the Associate Editor for Electronics! We look forward to receiving your next manuscript(s) in our on-line submission platform: http://ojs.midem-drustvo.si/index.php/InfMIDEM Last but not least, this issue brings 2 review and 5 original scientific papers. Enjoy reading them! Prof. Marko Topič Editor-in-Chief P.S. All papers published in Informacije MIDEM –Journal of Microelectronics, Electronics Components and Materials (since 1986) can be access electronically for free at http://midem-drustvo.si/journal/home.aspx. A search engine is provided to use it as a valuable resource for referencing previous published work and to give credit to the results achieved from other groups. Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 113 – 119 Micromachining of All-Fiber Photonic Micro-Structures for Microfluidic Applications Simon Pevec1, Borut Lenardič2 and Denis Đonlagić1 1University of Maribor, Faculty of Electrical Engineering and Computer Science, Maribor, Slovenia 2Optacore d.o.o., Ljubljana, Slovenia Abstract: Maskless micromachining of all-fiber photonics’ structures, based on the selective etching of structure forming optical fibers (SFF) is presented. A maskless micromachining process can reform or reshape a section of an optical fiber into a complex 3D photonic microstructure. This proposed micromachining process is based on the introduction of phosphorus pentoxide (P2O5) into silica glass through standard fiber manufacturing technology. Micro-machining is presented as a highly effective tool for the realization of new solutions in the design of optical sensors and microfluidic devices. Keywords: Micromachining; optical fibers; optical sensors; phosphorus pentoxide; selective etching; microstructures. Mikroobdelava povsem vlakenskih fotonskih mikrostruktur za področje mikrofluidnih aplikacij Izvleček: Predstavljena je mikroobdelava na osnovi selektivnega jedkanja posebnega optičnega vlakna v namene izdelave mikrofotonskih struktur. Proces mikroobdelave brez maskiranja sloni na vgradnji fosforjevega pentoksida (P2O5) v steklena optična vlakna skozi postopek standardne proizvodnje optičnih vlaken. Mikroobdelava je predstavljena kot zelo učinkovito orodje za realizacijo novih rešitev na področju načrtovanja senzorjev in mikrofluidnih naprav. Ključne besede: Mikroobdelava; optična vlakna; optični senzorji; fosforjev pentoksid; selektivno jedkanje; mikrostrukture. * Corresponding Author’s e-mail: simon.pevec@um.si 1 Introduction Photonic micro-structures are being increasingly used in a number of applications, ranging from optical tel­ecommunications [1-3] to biomedical sensor [4-6]. Direct implementation of micro-devices can extend their use and allow many important advantages and novel functionality. Existing solutions for optical fiber micromachining are mostly based on laser techniques. Reshaping of SiO2 optical fiber is typically realized by laser sources operating in ultraviolet (UV) or middle-infrared (MIR), where they have sufficient absorption of light in the SiO2 [7]. UV excimer lasers [8, 9] and fem­tosecond lasers [10-14] have been successfully applied as efficient tool for optical fiber micromachining. How­ever all direct laser techniques have a common need for individual and precision guiding of the laser beam over cylindrical optical fiber, which is complex, expen­sive and time-consuming task. Beside laser techniques, there are also other techniques like lithographic pro­cesses [15, 16], micromachining by dry etching [17] and by focused ion beam [18-21]. Lithographic process re­quires many process steps, dry etching is related with time-consuming and low selectivity etching, focused ion beam technique is also time-consuming and cost-inefficient and as such unsuitable for production. This paper presents a cost effective, mask-less mi­cromachining process that can re-shape a section of an optical fiber into a complex 3D photonic microstruc­ture. Micromachining based on selective etching pro­vides a unique way for efficient design and production of complex 3D photonic all-fiber microstructures and devices [22-25]. The selective chemical etching utilizes a phenomenon where the introduction of dopants into silica glass affects the etching rate of the glass when exposed to etching solution (usually HF). When pur­posely designed and properly doped optical fibers are combined/fusion spliced with standard fibers, selec­tive etching can be exploited for the manufacturing of micro-structures on the tip, along or within the optical fibers. Structures produced by this process are made entirely of silica glass, do not utilize any adhesives or foreign materials, and can thus sustain harsh chemi­cal and temperature conditions. Thus selective etching based micromachining involves production of SFF that involves preform production, mechanical reshaping of the preform and fiber drawing, fusion-splicing of these fibers with standard fibers, and etching of such assem­blies into final photonics microstructures or devices. Thus after proper SFF production, the device produc­tion is accomplished by a sequence of fiber cleave and splice sequence(s) that are followed by (wet) etching. This proposed micromachining process is mainly based on the introduction of P2O5 into silica, which can be ef­fectively removed upon exposing the fiber to the etch­ing medium. These preferentially etchable P2O5 doped areas within the fiber cross section can thus serve as sacrificial layers, thus allowing for the economical crea­tion of complex all-fiber devices, which will be present­ed as result of proposed micromachining technology. 2 Etching solutions and etching selectivity Etching selectivity S of doped region is defined as the ratio between the etching rate of the doped (vxx), and that of the pure silica (vSiO2). (1) The S depends on the dopant type, dopant concen­tration, etching medium and temperature. During in­vestigation several differently doped fiber preforms contained between one and five different doped layers of P2O5 concentrations were produced to study the im­pact of P2O5. Preforms with known refractive index profiles (one typ­ical refractive index profile is shown in Fig. 1) were in first step cut into approximately 1 to 2 cm long samples using a low speed diamond saw. The samples were then etched in etching medium. Depending on the compo­sition of the doped glass, an etching time of between 1 min and 3 h was used to obtain well-defined surface re­lief. The etching vessel was temperature stabilized and also vibrated to provide acid-mixing and the removal of etching by-products from the sample’s surface. The etched-preform samples were then cut in the axial di­rection through their centers, and were then analyzed/measured under an optical microscope. An example of such an etched preform measurement is shown in Fig. 2. The initial preform diameter and dimensions of the removed doped region were then used to deter­mine the average vxx/vSiO2 ratio of the individual layers of etched preform. Figure 2: Selectivity measurement of P2O5 doped pre­form obtained by comparison of geometrical shape of preform before and after etching. The preform analyzer data and the etching data were then combined to obtain a relationship between the etching selectivity S and the refractive index change caused by doping, which was further correlated to the dopant molar concentration [26]. Some dopants strongly increase S, while the others provide limited effect on the S at comparable concen­tration levels. From all researched dopants, P2O5 proved to be of particular interest for fiber micromachining. As shown in Fig. 3, the P2O5 doping of silica can provide very high etching selectivity S, even at low P2O5 con­centrations. Composition of an etching agent can also strongly influence the etching selectivity. Other do­pants can provide other benefits, such as strong index increase whilst providing very limited impact on the S (e.g. TiO2). Figure 3: Selectivity as a function of P2O5 and GeO2 in 40 % hydrofluoric acid at 25°C. Great impact on selectivity has also temperature of hy­drofluoric acid (HF), where by reducing the tempera­ture from 40 to -25 °C, selectivity of P2O5 doped sam­ple with higher dopant concentration (7.9 mol %) is increased from 28 to 39 as shown in Fig. 4. Figure 4: Selectivity as a function of temperature for two different P2O5 doped preform, etched in 40 % HF. Reducing the temperature has also negative impact on etching process, because it significantly slows down the absolute etching rate and thus increase the time required for the formation of the microstructure. Fur­thermore by adding isopropyl-alcohol (IPA) to HF, IPA-HF etching solutions work particularly well in combina­tion with P2O5 doping where doubling or even tripling of the etching selectivity can be achieved. 3 Micromachining of all-fiber photonics devices Fiber devices are created by splicing short section of SFF at the end-of lead or in-between two lead fibers. One, very simple example produced by selective etch­ing based on P2O5 doping is shown in Fig. 6. Here the micro-resonator on the optical fiber tip is presented. Micro-resonators have found applications within vari­ous photonic systems [27] such as sensors, filters, cou­pling devices, etc., but are difficult to produce. A cross section of the SFF, used for creation of resonator, is shown in Fig. 5 and consists of pure silica core, a large P2O5 region (5.7 mol%), and a thin pure SiO2 outer-layer with the same glass transition temperature as the lead-in fiber-cladding and, thus, allows for straightforward splicing between them. In this case the SFF was spliced between two coreless fibers, where the second coreless fiber was shortened to a length of about 15 mm as shown in Fig. 6b and then etched for sufficient time in HF. Figure 5: Optical microscopic cross-sectional view of SFF intended for micro-structure formation. The etchant first uniformly etched the entire structure, but once the pure silica outer-layer of the SFF was re­moved and HF came into contact with the P2O5-doped region, it preferentially removed this region entirely, leaving behind the final structure shown in Fig. 6a. The total etching time was 12 min in 40 % HF at 25 °C. Figure 6: (a) Scanning electron microscope view of the produced micro-resonator, (b) Fiber structure before etching (after cleaving and splicing). Depending of the device design, etching of SFF can be performed before or after splicing. Below are given few more typical examples of structures produced by ap­plication of splicing, cleaving and etching of SFFs. The first device shown in Fig. 7a is an all-fiber optical microcell that allows for the direct insertion of liquids, gases or solids within the optical path of the transmis­sion fiber. The micro-cell can be used as a transmission cell or as a miniature Fabry-Perot resonator. The total transmission loss of the microcell in Fig. 7a was less than 1 dB at 1550 nm, when immersed in water. Various lengths of micro-cells can be produced ranging from few tenths to few 1000 mm. Another example device that can be effectively pro­duced by this method is shown in Fig. 7b, and presents a miniature, all-silica, dual-parameter Fabry-Perot sen­sor for simultaneous measurement of surrounding fluid’s refractive index (RI) and temperature. This sensor permits a full temperature-compensated high resolu­tion RI measurement in range of 10-7 RIU, that can be used to determine very small changes in fluid structure or composition. All-silica design provides high chemi­cal and thermal inertness, while the miniature size pro­vides opportunities for measuring very small (nL) fluid volumes. Figure 7: Microstructure devices: (a) Microcell [28], (b) Refractive index – temperature sensor [29]. Next example shown in Fig. 8a presents an all glass, Fabry-Perot, fiber–optic pressure sensor. It is the world’s smallest commercial available pressure sensor [30], with outer diameter less than 125 mm, and is produced by proposed technology in few sequential steps on the tip of multi-mode lead-in optical fiber. Membrane thick­ness for typical pressure sensor is round 2 mm, which allow high pressure sensitivity needed for medical performance requirements. A sensitivity of 1100 nm/bar was also achieved which is, to our knowledge, the highest all-glass miniature sensor sensitivity reported in the literature. This robust sensor also demonstrated very high resistance to overload, which is an important advantage for practical usage of the sensor in realistic applications. The proposed miniature all-glass pressure sensor design is, therefore, a good candidate for appli­cations where size, cost, material inertness, mechani­cal and chemical resistance as well as insensitivity to electromagnetic interferences are important concerns. Beside all advantages coming from all silica glass opti­cal fiber design, this sensor can achieve high resolution and repeatability, very low drift, and fast response time. Figure 8: Microstructure devices: (a) Pressure sensor [31], (b1) SEM photo and (b2) optical microscope photo of Pressure – refractive index sensor respectively [32]. Another example shows one of more complex devices that can be produced by proposed technology, where microcell with pressure sensor was joined in series; it is multi parameter (multi cavity) Fabry-Perot sensor for simultaneous measurements of pressure and refractive index. Figure 8 (b1) shows scanning microscope (SEM) image, and Fig. 8 (b2) shows the same sensor under an optical microscope. These sensor was created at the tip of an optical fiber with a diameter that is equal to the standard fiber diameter, and length that does not exceeded 600 µm. High measurement resolutions bet­ter than 0.1 mBar and 2x10^-5 RIU can be achieved by using spectral interrogation and a FT-based measure­ment algorithm. Next example in Fig. 9a shows nanowire-based refrac­tive index sensor created on the tip of a single mode optical fiber configured as Fabry-Perot interferometer. Proposed micromachining technique including taper­ing allows creation of fiber coupled silica nanowires with radius between 200 and 600 nm. Nanowire sen­sor is made entirely of silica and includes a mechanical structure that provides stable operation and easy han­dling and packaging. High measuring spectral sensitiv­ity as 800 nm/RIU and low temperature sensitivity in water are typical sensor characteristics. Sensor might be an especially attractive platform for use in com­pact biochemical sensors, which utilize active surfaces, for example in various label-free detection-sensing schemes. One more example presented in Fig. 9b shows Fabry-Perot strain sensor created on the tip of a standard mul­ti-mode fiber. Sensor’s great advantage is simplicity of its production process that includes production of SFF (inset on Fig. 9b), which is cleaved, etched, and spliced between lead fibers in order to form final sensor. Tested sensors were successfully applied to strain-measure­ments exceeding 3000 me, which accommodate most of requirements encountered in practical industrial applications. A strain-resolution of 0.5 µ., high tem­perature range exceeding 650 °C, and low temperature intrinsic sensitivity below 0.04 nm/°C are typical char­acteristics for that kind of sensor. The last device shown in row is miniature all-silica fiber-optic sensor for simultaneous measurements of rela­tive humidity (RH) and temperature. The sensor is com­posed of two cascaded Fabry-Perot interferometers (FPIs) as shown in Fig. 9c. The first FPI consists of a short silica micro-wire (diameter is cca. 13 mm) coated by a thin layer of porous silica, and forms a RH sensing part. The second section created on the sensor tip forms a temperature measuring part. The typical total length of produced sensor is less than 2 mm, while diameter doesn’t exceed 125 mm. The sensor has good dynamic performances (rise time in few second range), it cover broad RH measuring range (0-100 %RH), and has linear characteristics for both measurement parameters with sensitivity of 0.48 degree/%RH and 3.7 degree/°C. All sensors and devices have all glass structure, high en­vironmental robustness, and a miniature design, which in any of the cases does not exceed the diameter of a standard optical fiber (i.e. 125 mm) and has an active length of less than 1.5 mm (more details can be found in appropriate references). All devices are robust and allow easy handling and packaging, especially those designed and fabricated on the tip of the optical fiber. Small dimensions, chemical resistance and robustness make sensors suitable for microfluidic applications. Since a single customized SFF production may result in the manufacturing of a large number of devices, the proposed process potentially presents a versatile and cost-efficient way of producing all-fiber devices or de­vice sub-assemblies. 4 Conclusions An effective technique for production of all fiber de­vices through application of selective etching and spe­cially designed SFF was presented. 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Donlagic, “Miniature fiber-optic sensor for simultaneous measurement of pres­sure and refractive index,” Opt Lett 39, 6221-6224 (2014). 33. S. Pevec and D. Donlagic, “Nanowire-based re­fractive index sensor on the tip of an optical fiber,” Appl Phys Lett 102(2013). 34. S. Pevec and D. Donlagic, “All-fiber, long-active-length Fabry-Perot strain sensor,” Opt Express 19, 15641-15651 (2011). 35. S. Pevec and D. Donlagic, “Miniature all-silica fib­er-optic sensor for simultaneous measurement of relative humidity and temperature,” Opt Lett 40, 5646-5649 (2015). Arrived: 31. 08. 2016 Accepted: 22. 09. 2016 S. Pevec et al; Informacije Midem, Vol. 46, No. 3(2016), 113 – 119 Figure 1: Preform analyzer data obtained after MCVD preform production. S. Pevec et al; Informacije Midem, Vol. 46, No. 3(2016), 113 – 119 S. Pevec et al; Informacije Midem, Vol. 46, No. 3(2016), 113 – 119 S. Pevec et al; Informacije Midem, Vol. 46, No. 3(2016), 113 – 119 S. Pevec et al; Informacije Midem, Vol. 46, No. 3(2016), 113 – 119 Figure 9: Microstructure devices: (a) Nano-wire refrac­tive index sensor [33], (b) Strain sensor [34], and (c) Relative humidity – temperature sensor [35]. S. Pevec et al; Informacije Midem, Vol. 46, No. 3(2016), 113 – 119 Review scientific paper Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 120 – 129 Design and analysis of differential passive circuits for I/Q generation in 60 GHz integrated circuits Ivan M. Milosavljević1,2, Dušan P. Krčum1,2, Lazar V. Saranovac1 1Department of Electronics, School of Electrical Engineering, University of Belgrade, Belgrade, Serbia 2NovelIC Microsystems, Belgrade, Serbia Abstract: I/Q generation circuit design is of great importance within the design of modern radars integrated on chip, as well as in high-speed communication circuits operating at millimeter wave frequencies. Due to increased robustness on process variations and noise, differential signaling and differential I/Q generators are of particular interest. Several passive topologies suitable for usage in the commercially available 130 nm SiGe BiCMOS process are presented and evaluated. These topologies are: branch-line coupler (BLC), broadside coupler (BSC), poly-phase filter (PPF) and quadrature all-pass filter (QAF). The first three topologies are implemented and the obtained results are compared to results published previously. The designed PPF has a total phase error of ± 2.5° over a 40 GHz bandwidth and this is the most desired solution if the total available area is a limiting factor. Transmission lines required for the design of BLC and BSC are small enough, making such structures easy to implement using today’s mainstream technologies. BLC is a reliable and widely used solution for I/Q generation at almost any microwave frequency. Designed BLC has a phase error of ± 1.7° over a 7 GHz bandwidth. BSC has proved to be the best solution for I/Q generation in the 60 GHz band. The designed solution has a smaller area than BLC, a phase error of only ± 5° over a 40 GHz bandwidth and ± 1° over a 7 GHz bandwidth. Keywords: millimeter-wave passive circuits; I/Q generation; 90o hybrid coupler; branch-line coupler; broadside coupler; poly-phase filter; quadrature all-pass filter Načrtovanje in analiza diferencialnih pasivnih vezij za generiranje I/Q v 60 GHz integriranih vezjih Izvleček: Načrtovanje I/Q generirnih vezij ime velik pomen v modernih integriranih radarjih, kakor tudi pri komunikacijskih vezjih visokih hitrostih v območji milimetrskih valovnih dolžin. Zaradi proizvodne robustnosti in šuma, so difencialni I/Q generatorji zelo pomembni. Predstavljene so številne pasivne topologije uporabne v komercialni 130 nm SiGe BiCMOS tehnologiji, in sicer: linijski sklopnik (BLC), hibridni sklopnik (BSC), polifazni filter (PPF) in kvadraturni polnoprepustni filter (QAF). Prve tri topologije so uporabljene in primerjane s prej objavljenimi rezultati. Načrtovan PPF ima skupno fazno napako ±2.5° pri 40 GHz pasovni širini in je najugodnejša rešitev, če imamo omejitve s prostorom. Prenosne linije, ki so potrebne za BLC in BSC so dovolj majne, da lahko ti topologiji uporabimo z uporabo današnjih tehnologij. BLC je zanesljiva in najbolj uporabljena rešitev v mikrovalovnih frekvencah. Fazna napaka načrtanega BLC je ±1.7° v 7 GHz pasovni širini. Najugodnejša rešitev v 60 GHz pasu je BSC. Potrebuje manjšo površino in ima fazno napako ±5° v 40 GHz pasu in ±1° v 7 GHz pasu. Ključne besede: milimeterska valovna pasivna vezja; I/Q generacija; 90° hibridni sklopnik; vejni sklopnik; polifazni filter; kvadraturni polnoprepusti filter * Corresponding Author’s e-mail: ivan.milosavljevic@novelic.com © MIDEM Society I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 1 Introduction Today’s millimeter-wave integrated circuits are pri­marily intended for communication and radar sensing applications. Advancements in silicon-based technol­ogy processes allowed the very high scale integration, making complete millimeter-wave systems integrated into a single chip. In these complex systems, usually there is a need for the generation of in-phase (I) and quadrature (Q) signals [1-5]. As an example, for blocks such as I/Q modulators [6], frequency multipliers [7], or vector-modulators [8-10], it is mandatory to have supporting circuits that provide the desired 90o phase shift between the input signals, in cases when quad­rature signals are not provided by the local oscillator (LO). Direct I/Q signal generation from the LO requires the design of a quadrature voltage controlled oscil­lator (QVCO). Although the design of a conventional parallel-coupled QVCO at millimeter-wave frequencies is feasible and reported in [11], the approach suffers from very high phase noise, and therefore its usage is limited. There are other QVCO state-of-the-art tech­niques based on injection- [12] and magnetic-coupling [13] that improve phase-noise, but their complexity and reliability issues rise with frequency. At microwave frequencies 90o phase generation can be provided by active quadrature dividers [14]. The use of dividers requires the VCO operating at twice the nominal fre­quency, therefore at higher millimeter-wave frequen­cies this is not a suitable topology. Another approach is using injection-locked frequency multipliers [15], that also suffer from similar problems as the approach with QVCO. Furthermore, using active circuits for I/Q gener­ation requires current-hungry and more sophisticated designs at millimeter-wave frequencies. In opposite to active circuits, passive circuits do not consume power. However, they introduce some losses that often need to be compensated, and the total loss per branch can­not be smaller than the theoretical minimum of 3 dB. Besides consumption, the key advantages of passive circuits are design simplicity and reliability. Conse­quently, a natural choice for I/Q signal generation at millimeter-wave frequencies is a passive circuit. There are various ways of generating quadrature sig­nals with passive circuits based on quarter-wave (./4) coupled and transmission line millimeter-wave struc­tures. The limiting factor for quarter-wave structures, also known in literature as 90o hybrid couplers, is chip area which is directly related to signal wavelength. This is the reason why these structures are much more in­teresting nowadays when the majority of high-speed communication systems operating frequencies are shifted to the millimeter-wave area. Passive quarter-wave structures for microwave frequencies have been investigated at various substrates. The investigations are mainly focused on miniaturization of quarter-wave coupled structures, as reported in [16, 17]. Most of them only deal with single-ended circuits, not suitable for integration in high-performance integrated circuits. Therefore this paper is focused only on topologies that have differential counterparts, in order to minimize sensitivity to noise and undesired couplings. On the other hand, passive circuits for quadrature generation are also designed from lumped components. The main representatives are RC and RLC based circuits and a vast number of papers comprised of circuits at millime­ter-wave frequencies have been reported [18-20]. The article is organized as follows: Section 2 shows the­oretical basics of different passive circuits for I/Q gener­ation. Implementation of circuits and simulation results obtained with electromagnetic (EM) solvers are shown in Section 3, followed by conclusions in Section 4. 2 Passive circuits for I/Q generation The main parameters of passive circuits for I/Q genera­tion are reflection coefficients, insertion loss, coupling, directivity, bandwidth, phase imbalance, amplitude imbalance, isolation between output ports and chip area. The priority of parameters is determined by spe­cific application. Typically in millimeter-wave integrat­ed circuits, the highest priority parameters are phase imbalance, insertion loss and chip area. Based on physical phenomena of quadrature genera­tion, passive circuits are divided in two main categories: I/Q generators based on distributed structures (trans­mission lines) and I/Q generators based on lumped components. 2.1 Distributed Transmission line-based passive circuits provide 90o phase shift using quarter-wave segments of transmis­sion lines. The most basic representative is the one with ./4 transmission line inserted in the quadrature signal path. This approach requires a power divider for splitting the input signal in two signals of equal power. If a Wilkinson power divider [21] is used, sig­nificant chip area is required. In Figure 1, differential structures based on 90o transmission lines are shown. The input differential signal is LO, and the outputs are differential in-phase (I) and quadrature (Q) signals. Con­ventional approach shown in Figure 1 (a) suffers from narrowband operation, which can be improved using the Schiffman technique shown in Figure 1 (b) [22]. With this technique, performance is improved at the expense of chip area. Both implementations are im­practical in millimeter-wave integrated circuits due to increased area and attenuation. Alternatively, distributed structures based on concen­trated coupling and coupled transmission lines are widely used in millimeter-wave integrated circuits. Figure 1: (a) 90o transmission line based power divider, (b) Schiffman 90o power divider. 2.1.1 Concentrated coupling Branch-line coupler (BLC) is a representative of trans­mission lines concentrated coupling. Conventional BLC hybrid is shown in Figure 2 (a). Since it occupies consid­erable chip area, reduction of transmission lines length is proposed in [16], by inserting shunt capacitance at the end of the transmission lines. This is shown in Fig­ure 2 (b). Figure 2: (a) Conventional branch-line coupler, (b) Re­duced size branch-line coupler. The relationships between electrical lengths of branch-line q1 and through-line q2, the differential character­istic impedance Z, the nominal differential impedance Zdiff, and shunt capacitance C in reduced-size BLC hy­brid are given as (1) (2) . (3) In Figure 2 (b) it is given commonly used case when Zdiff /Z=1/.2, q1 = 30o and q2 = 45o. These values are very suited for practical implementations. Additionally, a technique of meandering is implemented whenever applicable to reduce total chip area. 2.1.2 Coupled line In microwave theory, I/Q generators based on coupled lines are known as directional couplers. Directional couplers are easily constructed from two ./4 coupled transmission lines. The coupling effect is achieved by broadside coupling or edge-side coupling, as shown in Figure 3. The coupling ratio between two broadside coupled metal layers is given by (4) Where Z0e and Z0o are even- and odd-mode impedances of the coupled lines, er is the relative permittivity of the substrate, and kair is the coupling factor of the two lines in air. Figure 3: (a) Broadside coupling, (b) Edge-side cou­pling. In integrated realizations, the broadside coupler (BSC) is suitable for implementation, and its vertical ap­proach saves chip area. An important issue is that the distributed capacitance from the broadside coupled lines to the ground is asymmetric. Therefore, the width of transmission lines must be asymmetric to minimize asymmetry and optimize bandwidth, as shown in Fig­ure 3 (a). Alternatively, edge-side coupled structures shown in Figure 3 (b) can be used. They are very simple to de­sign, but suffer from several issues. Due to its inhomo­geneous dielectric nature, modal velocities of even- and odd-modes are different. This difference leads to poor isolation and, consequently, worse directivity of the coupler. In order to overcome this problem, modal velocities should be equalized. Within practical solu­tions for equalization, the most suitable one for inte­grated circuit (IC) design is the placement of capacitors between input and output ports. However, even after applying this technique, overall performance of edge-side coupler remains mediocre. The edge-side coupler also suffers from reliability issues due to fabrication and tolerance problems. As a solution, multiple transmission lines are interdigi­tated in edge-side couplers. This structure is known as a Lange coupler [23], and the key design parameter is the voltage coupling coefficient c. Improved coupling helps in relaxing fabrication and tolerance problems. Millimeter-wave designs of single-ended unfolded Lange coupler are reported in [8, 24], etc. Differential Lange coupler requires two single-ended Lange struc­tures that significantly increase the area and complex­ity of the circuit, therefore they are rarely used. 2.2 Lumped components based Passive circuits for quadrature signal generation can be realized using basic circuit elements such as resistors, capacitors and inductors. Mentioned elements can eas­ily be implemented in currently available silicon-based processes. This approach in generation of quadrature signals leads to simple, inexpensive and very compact design. However, this design has large losses and a sig­nificant central frequency shift which is temperature and process dependent. Circuits for I/Q generation that use lumped elements are unable to handle larger pow­er levels. In practical designs, two main circuit topolo­gies are used: poly-phase filter (PPF) and quadrature all-pass filter (QAF). 2.2.1 Poly-phase filters PPF quadrature generation is based on phase shaping using combination of low-pass and high-pass RC filters. Low-pass (LPF) and high-pass filter (HPF) transfer func­tions are given by (5) (6) Corresponding arguments are: (7) (8) If the values of the components in LPF and HPF are cho­sen as , where w0 is radial frequency of an input signal, phase shift through a LPF is - 45o and through HPF is + 45o, while attenuation is equal in both paths. The first order PPF achieves a narrowband quadrature generation and has a wide phase variation over process and temperature. More robust design can be obtained using higher order PPF. Second order PPFs are widely used, as shown in Figure 4. In order to reduce influence of the process variations, component values of each PPF stage are chosen for a different central frequency: (9) (10) Relationship between the input radial frequency and poles of PPF is given by (11) where . Figure 4: Second order PPF schematic. 2.2.2 Quadrature all-pass filter QAF is used for phase shaping of quadrature output signals. In Figure 5, is shown the schematic of a fully differential QAF. Theory of operation is similar to PPF. Namely, QAF also uses combination of LPF and HPF for adequate phase shift between the output signals. Transfer functions be­tween LO input and I and Q outputs are given in equa­tions (12) and (13): (12) (13) Figure 5: LC based QAF schematic. Phase difference between QAF outputs can be derived from the previous transfer functions, and after simplifi­cation the final expression for the QAF phase difference is: (14) In order to make the phase difference between I and Q signals equal to 90o, the resistance value should be (15) and for equal attenuation in both paths (16) 3 Implementation and simulation results In this section, different passive I/Q generators are im­plemented using a commercially available 130 nm SiGe BiCMOS technology with seven metal layers shown in Figure 6. In all implementations solid metal 1 is used as circuit ground. All passive circuits are simulated in 2.5D planar EM solver based on method of moments (MoM), and verified in 3D full-wave solver based on finite element method (FEM). The comparison between simulation results is given for all designed circuits. The results ob­tained with MoM based simulator are shown as dashed lines, and the results obtained with FEM based simula­tor as solid lines. Impact of the mismatch and process variations of resistors and capacitors on phase and amplitude imbalance is simulated for all circuits using Monte Carlo simulations. Influence of the output load mismatch is also analyzed. Output impedances of both I and Q loads are varied in range 80% - 120% of nominal 100 . differential impedance. Effect of the mismatch and process variations and load mismatch is analyzed on single frequency of 60 GHz and characterized using ±3. (standard deviation). 3.1 Branch-line coupler Implementation of differential reduced-size BLC is presented in this subsection. Transmission lines are implemented in 3 mm thick top metal 2. Differential impedance of branch-line and through-line is 141.4 ., which corresponds to lines of width 3 mm and spacing 17 mm. Transformation to a differential impedance of 100 . at input and output ports is achieved by reduc­ing line spacing to 7 mm and retaining same widths. According to equation (3), 43 fF differential capacitor value is obtained. Metal-insulator-metal (MIM) capaci­tors are used. In this technology, the bottom electrode of the MIM capacitor is connected to metal 5 and the top electrode to top metal 1. Thus, two serial capacitors with twice the capacitance are used instead of a sin­gle differential capacitor. Capacitors are made in such a way to fit between differential lines and are connected in metal 5. 3D preview of the reduced-size BLC is shown in Figure 7. A meandering technique is used to reduce the overall BLC area, which is 267 mm x 230 mm. The termination port is connected to a 100 . resistor and the performance of the 60 GHz BLC obtained by MoM and FEM based simulators are shown in Figure 8. The BLC exhibits excellent performance in a 7 GHz bandwidth. Phase imbalance of the BLC is lower than 1.7o, and amplitude imbalance is lower than 0.4 dB. Reflection coefficients at input and outputs are lower than -14 dB. Performing process and mismatch Monte Carlo simulations gave us the insight in circuit’s be­havior under mentioned conditions. It is noticed that phase and amplitude imbalance does not vary more than 2.42o and 0.14 dB in the case of process and mis­match variations of the termination resistor and MIM capacitors. Major contributor on phase and amplitude imbalance deviation is the process variation of MIM capacitors and this should be taken into account in design process of reduced-size BLC. I/Q load mismatch effect on the phase imbalance is lower than 0.76o, while the amplitude imbalance is negligible. 3.2 Broadside coupler Differential input and output signals of the BSC are im­plemented in top metal 2 layer. Transmission lines are optimized to a differential impedance of 100 .. The width of lines is 3.5 mm and spacing is 4.5 mm. Coupled ./4 lines are implemented in top metal 1 and metal 5 to achieve high coupling efficiency. Inter-layer distance is 0.85 mm. It requires the coupling coefficient k of 1/.2, and according to equation (4) values for even- and odd-mode impedances are approximately 241.5 . and 41.4 .. The widths of coupled lines are calculated from the EM simulation and top metal width of 5 mm and bottom metal width of 8 mm are obtained. In Figure 9, the geometry of 60 GHz differential BSC is shown. Ag­gressive meandering of the coupled lines is performed to minimize chip area. The BSC dimensions are 190 mm x 170 mm. Termination impedances T1 and T2 are chosen to be asymmetric in order to achieve very low phase mis­match. In this way the amplitude difference is con­sciously sacrificed. Termination impedance T1 is 20 ., and termination impedance T2 is 50 .. The perfor­mance of the 60 GHz BSC obtained by MoM and FEM simulators are shown in Figure 10. In the 40 GHz bandwidth, phase imbalance of the BSC is lower than 5o, and amplitude imbalance is lower than 2 dB. Reflection coefficients at inputs and outputs are lower than -10 dB, thus an excellent matching is achieved without any additional matching structures. Impact of the mismatch and process variations of ter­mination resistors on phase and amplitude imbalance is lower than 1.1o and 0.18 dB. Also, the load mismatch effect on the phase and amplitude imbalance is lower than 1.5o and 0.25 dB. 3.3 Poly-phase filter Two stage PPF is designed for 60 GHz band as fol­lows. Central operating frequency f1 for the first stage is 62.25 GHz, and for the second stage f2 is 59.25 GHz. Capacitance is chosen to be the same in both stages and equal to 39 fF. This is quite small value for the given technology, thus the series connection of four capaci­tors with capacitance of 156 fF is used. The resistance of the first stage is then R1 = 64.75., and the resistance of the second stage is R2 = 68.. Dimensions of the designed PPF are 130 mm x 80 mm, and 3D preview is shown in Figure 11. The performance of the 60 GHz PPF are shown in Fig­ure 12. The phase imbalance is less than 2.5o at a 40 GHz range, but the PPF has significantly poorer matching. Impact of the mismatch and process variations of resis­tors and MIM capacitors on the phase and amplitude imbalance is lower than 0.6o and 0.1 dB making PPF very robust to mismatch and process variations. Load mismatch effect on the phase and amplitude imbal­ance is lower than 3.8o and 1 dB, which shows that the PPF is very sensitivity on output load mismatch. The performance summary is shown in Table 1. 4 Conclusion An overview of I/Q generation circuits for the 60 GHz band is presented in this paper. The three most suit­able and most common topologies are chosen to be designed in a modern BiCMOS process. In comparison to other I/Q generation circuits, BSC ex­hibits the best performance. The main advantages of this design over other references is a small area and small phase error over a 40 GHz frequency range. In­sertion loss is comparable to theoretical values, thus it does not require an additional amplification stage. In a 7 GHz bandwidth around 60 GHz it presents supreme performance. All of these parameters highlight BSC as the first choice for an I/Q generation block in modern CMOS designs, in millimeter-wave integrated circuits. A possible drawback of this design is the usage of a specific metal stack. Depending on the vertical spacing between metal layers used for the implementation of the coupler, desired even- and odd-mode impedances may not be achievable. On the other hand, BLC is implemented in the same metal layer and does not have the technology related drawbacks. Small areas can be obtained using mean­dering and capacitive termination of ./4 lines. How­ever, using capacitive termination of ./4 lines leads to increased phase and amplitude deviation related with process and mismatch variations of used capacitors. Phase error in a 7 GHz bandwidth around 60 GHz is no­ticeably low and all ports are well matched. Insertion loss and output isolation are equally good as for BSC. However, BLC occupies a larger area than BSC. A poly-phase filter is used as the third I/Q generation circuit for applications in the 60 GHz band. This solution can be used in situations where area is critical. Howev­er, PPF has huge insertion losses and has large sensi­tivity on the output load impedance mismatch. These are the reasons why PPF requires additional output buffer amplifiers. PPF matching is also poor, and ad­ditional matching networks are mandatory. The great­est advantage of PPF is its robustness on process and mismatch variations, and a very small phase difference over a very wide frequency range. 5 References 1. C. Marcu, D. Chowdhury, C. Thakkar, J.D. Park, L.K.Kong, M. Tabesh, Y. Wang, B. Afshar, A. Gupta, A. Arbabian, S. Gambini, R. Zamani, E. Alon, and A.M. 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Hy­brid Coupler Using Coupled-Line Section With Spurious Rejection,” IEEE Microwave and Wireless Components Letters, vol. 24, no. 11, pp. 766–768, November 2014. 18. M.G.M. Notten and H. Veenstra, “60GHz Quadra­ture Signal Generation with a Single Phase VCO and Polyphase Filter in a 0.25µm SiGe BiCMOS technology,” in IEEE Bipolar/BiCMOS Circuits and Technology Meeting. 2008, pp. 178–181, IEEE. 19. I. Sarkas, M. Khanpour, A. Tomkins, P. Chevalier, P. Garcia, and S.P. Voinigescu, “W-band 65-nm CMOS and SiGe BiCMOS transmitter and receiver with lumped I-Q phase shifters,” in 2009. RFIC 2009. IEEE Radio Frequency Integrated Circuits Sympo­sium, Boston, 2009, pp. 441–444, IEEE. 20. W. Shin and G.M. Rebeiz, “60 GHz active phase shifter using an optimized quadrature all-pass network in 45nm CMOS,” in 2012 IEEE MTT-S In­ternational Microwave Symposium Digest (MTT), Canada, 2012, pp. 1–3, IEEE. 21. E.J. Wilkinson, “An N-Way Hybrid Power Divider,” IRE Transactions on Microwave Theory and Tech­niques, vol. 8, pp. 116–118, January 1960. 22. B.M. Schiffman, “A New Class of Broad-Band Mi- crowave 90-Degree Phase Shifters,” IRE Transac­tions on Microwave Theory and Techniques, vol. 6, pp. 232– 237, April 1958. 23. J. Lange, “Interdigitated Strip-Line Quadrature Hybrid,” in 1969 G-MTT International Microwave Symposium. 1969, pp. 10–13, IEEE. 24. M.K. Chirala and B.A. Floyd, “Millimeter-Wave Lange and Ring-Hybrid Couplers in a Silicon Tech­nology for E- Band Applications,” in 2006. IEEE MTT-S International Microwave Symposium Di­gest. 2006, pp. 1547–1550, IEEE. 25. B.A. Floyd, S.K. Reynolds, U.R. Pfeiffer, T. Zwick, T. Beukema, and B. Gaucher, “SiGe bipolar trans­ceiver circuits operating at 60 GHz,” IEEE Journal of Solid- State Circuits, vol. 40, pp. 156–167, Janu­ary 2005. Arrived: 17. 05. 2016 Accepted: 22. 08. 2 016 I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 Figure 6: Technology metal stack. Figure 7: 3D preview of the differential branch-line coupler. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 Figure 8: The performance of differential branch-line coupler: (a) phase difference, (b) insertion loss, (c) re­flection coefficient, (d) isolation. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 Figure 9: 3D preview of the differential broadside cou­pler. Figure 10: The performance of differential broadside coupler: (a) phase difference, (b) insertion loss, (c) re­flection coefficient, (d) isolation. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 Figure 11: 3D preview of the 60 GHz poly-phase filter. Figure 12: The performance of the 60 GHz poly-phase filter: (a) phase difference, (b) insertion loss, (c) reflec­tion coefficient, (d) isolation. I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 Table 1: Overview of 60 GHz passive circuits for I/Q generation. Reference Type Pol.a BW [GHz] |Sii|b [dB] |Sij|c [dB] |S32| [dB] PI [0] AI [dB] Area [mm2] [1] BLC S.e. 7 -11 -6 -14 ± 3.6 ± 0.75 370x270 [24] Lange S.e. 60 -16 -4.5 -14 ± 9.6 ± 2.25 300x160 [24] Lange S.e. 60 -13 -4.5 -11 ± 11.5 ± 1.75 160x120 [5] BSC S.e. 30 -15 -4.5 N/A ± 3 ± 0.75 280x200 [6] BSC Diff. 40 -13 -5 N/A ± 5.3 ±0.6 350x175 [25] BLC Diff. 7 N/A N/A N/A ± 4 ± 0.5 400x550 [18] PPF Diff. 2 N/A -10 N/A ± 2.1 N/A 210x200 This workd BLC Diff. 7 -14 -4.5 -17 ± 1.7 ± 0.4 267x230 This workd BSC Diff. 40 -10 -4.5 -15 ± 5 ± 1.5 190x170 This workd PPF Diff. 40 -5 -14 -22 ± 2.5 ± 0.75 130x80 a polarity (Pol.): single ended (S.e.) or differential (Diff.) b input and output reflection coefficients: worst or any known values c insertion loss: average or any known value d simulation results I. M. Milosavljević et al; Informacije Midem, Vol. 46, No. 3(2016), 120 – 129 Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 130 – 135 Dual-Mode Sinusoidal Quadrature Oscillator with Single CCCTA and Grounded Capacitors Worapong Tangsrirat Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang (KMITL), Ladkrabang, Bangkok, Thailand Abstract: In this work, a sinusoidal quadrature oscillator which simultaneously generates voltage and current signal outputs is proposed. It contains only a single current-controlled conveyor transconductance amplifier (CCCTA) and two grounded capacitors. The proposed oscillator has the advantage features of resistorless structure realization, electronic frequency control, availability of two explicit voltage and current quadrature outputs, and low sensitivity figure. Moreover, the parasitic elements existing at the CCCTA terminals are taken into account. The performance of the proposed oscillator circuit was verified using PSPICE simulation with acceptable results. Keywords: Current-Controlled Conveyor Transconductance Amplifier (CCCTA); Quadrature Oscillator; Resistorless circuits; Voltage-mode and current-mode circuits. Dvojni sinusni kvadrantni oscillator z enosnim CCCTA in ozemljenimi kondenzatorji Izvleček: V članku je predstavljen je kvadrantni oscilator, ki vzporedno generira napetostni in tokovni signal. Vsebuje le en tokovno krmiljeno vezje transkonduktančnega ojačevalnika in dva ozemljena kondenzatorja. Predlagano vezje je brez uporov, vsebuje elektronski nadzor frekvence, omogoča dva ločena napetostna in tokovna izhoda in izkazuje nizko občutljivost. Upoštevani so parazitni elementi na CCCTA terminalih. Lastnosti oscilatorja so bile preverjene v PSPICE okolju. Ključne besede: tokovno krmiljeno vezje transkonduktančnega ojačevalnika; brez uporovno vezje;napetostno in tokovno vezje * Corresponding Author’s e-mail: drworapong@gmail.com 1 Introduction Sinusoidal quadrature oscillator or two-phase sinusoi­dal oscillator is a kind of the sinusoidal oscillators that provides explicit two signal outputs with 90° phase shift from the same structure. Accordingly, it performs an essential circuit block employed a wide range of appli­cations in modern electronic and communication sys­tems, control systems, and signal processing. There are many attempts recently in designing sinusoidal quad­rature oscillators based on various types of modern ac­tive components [1-16]. However, many of them make use of at least two active components [1-12]. Only few circuits can provide both voltage and current quadra­ture signals from the same configuration [11-14]. These oscillator realizations contains an excessive number of external passive components, i.e., at least four passive components. The recent current-mode quadrature os­cillators based on single current differencing transcon­ductance amplifier (CDTA) were introduced in [15-16]. The previous work in [15] employs only one CDTA and three passive components (including two virtually grounded passive components that are floating in the non-ideal sense). In [16], a compact single CDTA-based quadrature oscillator with three external passive com­ponents was reported. This circuit requires a floating capacitor, which is not favorable for further integration. In 2008, the recently defined active circuit element, the so-called current-controlled conveyor transconduct­ance amplifier (CCCTA), was introduced [17]. This de­vice is a modified conception of the current conveyor transconductance amplifier (CCTA) [18], in which its parasitic resistance seen at the x-terminal (Rx) is vari­able electronically by adjusting an external biasing current. This property provides the advantage of realiz­ing electronically controllable analog function circuits without external passive resistor requirement. Since its introduction, the CCCTA has numerous applications in a class of analog signal processing solutions and cir­cuits [17], [19-21]. This paper presents a sinusoidal oscillator with variable oscillation frequency, able to provide explicitly quadra­ture voltage and current outputs from the same circuit configuration. The proposed quadrature oscillator em­ploys only one CCCTA and two grounded capacitors. A detailed analysis shows that the oscillator circuit in­cludes low active and passive sensitivities and has good frequency stability. Moreover, the effects of the CCCTA parasitic elements on the oscillator performance are also discussed. Simulation results with PSPICE using standard 0.35-mm BiCMOS process parameters are per­formed to verify the practical utility and validity of the realized circuit. 2 Principle of the CCCTA and its realization Basically, the CCCTA can be realized through a cascade connection of second generation current-controlled conveyor (CCCII) and multi-output transconductance amplifier. Fig.1 shows the electrical symbol and equiva­lent circuit of the CCCTA. It is shown that this device consists of two input terminals (y and x) and two out­put terminals (z and o±). An ideal property of the CC­CTA is described by the following matrix : (1) where Rx represents the parasitic serial resistance at the x-terminal, and gm denotes the effective small-signal transconductance gain of the CCCTA. As described in eq. (1), the x-terminal has a parasitic resistance Rx, where its value usually depends on an external sup­plied current. The y-terminal exhibits the high-input impedance terminal, while the z and o-terminals are two types of high-output impedance terminals. One possible realization of the CCCTA in BiCMOS technology is shown in Fig.2 [22]. The circuit is mainly composed of second-generation current-controlled conveyor (Q1-Q2, M1-M7) and dual-output transcon­ductance amplifier (Q3-Q6, M8-M14). Referring to Fig.2, the parasitic resistance Rx of the CCCTA has been de­rived as : (2) where VT is the thermal voltage, whose value is approxi­mately 26 mV at 27oC. Note from eq.(2) that the value of Rx depends on the external DC bias current IA. Assum­ing transistors Q3-Q5 as well as M8-M11 are matched, the expression of gm can be given by : (3) Also note that the gm-value is controllable electroni­cally and linearly by changing the IB–value. 3 Proposed dual-mode sinusoidal quadrature oscillator Fig.3 shows a canonic sinusoidal oscillator that produc­es voltage and current quadrature outputs explicitly. The circuit constructs from only one CCCTA and two grounded capacitors without needing any external passive resistor. The state-space equations for this con­figuration is obtained as [23]-[24] : (4) where (5) From the above autonomous state-space expression, the characteristic equation of the circuit can be derived as : (6) The condition of oscillation and the frequency of oscil­lation (wo) from eq.(6) are expressed, respectively, by (7) and (8) This means that the circuit will oscillate with no oscillation condition at the oscilla­tion frequency of (9) It is obvious that the wo is electronically tunable through the transconductance gain (gm) and/or parasitic resist­ance (Rx) of the CCCTA. Thus, the circuit can work as an electronically variable frequency quadrature oscillator. Considering the proposed configuration of Fig. 3, the two output voltages marked v1 and v2 are related as : (10) where k1 = woRxC2. Eq.(10) represents a 90°-phase differ­ence between both voltages, showing the quadrature property of the proposed oscillator. Furthermore, in case of k1 = 1, the amplitudes of two quadrature out­puts will also be equal. In addition, it is crucial to note that the quadrature output voltages v1 and v2 are not in low-impedance levels, hence external voltage buffers are necessary Also from Fig.3, the relation for two output currents (i1 and i2) can be given by the following matrix equation. (11) It is seen that, in this case, the relationship between two quadrature current outputs i1 and i2 can be obtain as : (12) where k2 = woC2/gm. Clearly, for k2 = 1, two marked ex­plicit quadrature current outputs have equal magni­tude. It is also to be noted that the circuit provides the output current i1 from the high-impedance terminal (terminal o+) but the output current i2 can be obtained across C2. Therefore, for explicit dual-mode utilization, an external buffering unit would be required for sens­ing and taking out the current i2. According to eq. (9), the relative sensitivity of wo with respect to active and passive components can be ob­tained as : (13) All of which are lower than unity in magnitude. 4 Effects of the CCCTA Parasitic Elements Fig.4 shows the practical model of the CCCTA. As it is seen, there are parasitic resistances and capacitances from terminals y, z and o± to the ground (Ry //Cy , Rz //Cz and Ro //Co), and a serial parasitic resistance Rx at the x-terminal. It is further to be noted that the typical values of parasitic resistances Ry, Rz and Ro are in the range of several MW, whereas parasitic capacitances Cy, Cz and Co are within a few fFs. Consider the CCCTA parasitic ele­ments in the proposed oscillator of Fig.3. It is clear that the external grounded capacitors C1 and C2 are parallel connected at the terminals y and z, respectively. The effects of parasitic capacitances at corresponding ter­minals could be adsorbed, as they merge with external capacitance values. Hence, the total impedance at the y-terminal can be approximated to : (14) For the working frequencies, (15) Zy can be further reduced to the value of 1/C1s, which is practically not affected by Ry//Ro. In a similar way, at the z-terminal, the influence of Rz can also be alleviated for operation at frequencies: (16) As a result, it can be realized from eqs. (15) and (16) that the frequency range at low frequencies should be se­lected as [25]: (17) Furthermore, it should be considered that there is a high-frequency limitation owing to the parasitic im­pedances (Ro //Co) in parallel at the terminal o+. Thus, the extra pole introduced at the terminal o+ can be expressed as : wo @ 1/(RoCo). To exhibit the ideal charac­teristic, the operating frequency range at high frequen­cies is found as : (18) Finally, combining eqs.(17) and (19), the useful frequen­cy range of the proposed oscillator can be defined as : (19) 5 Computer Simulation and Performance Verification The proposed dual-mode sinusoidal oscillator as de­picted in Fig.3 was simulated using PSPICE program. In simulation purpose, the CCCTA structure given in Fig.2 was employed with standard 0.35-mm BiCMOS pro­cess parameters using supply voltages of +V = -V = 1 V. The aspect ratios (W/L in mm/mm) of the MOS transis­tors were set to 7/0.7 and 8.5/0.7 for all the NMOS and PMOS transistors respectively. By choosing C1 = C2 = 0.4 nF, IA = IB = 25 mA, the proposed oscillator circuit of Fig.3 was designed to oscillate at fo = wo/2p @ 191 kHz. By performing time-domain analysis, the simulated transient waveforms for quadrature volt­age and current outputs of the proposed oscillator are shown in Figs.5 and 6, respectively. As obtained from simulation results, the frequency of oscillation (fo) was observed as 185 kHz. Fig.7 also shows the simulated fre­quency spectrums of both voltage and current quad­rature output waveforms, and the observed values of total harmonic distortion (THD) at all the outputs were less than 2.89%. To further demonstrate the electronic frequency controllability of the oscillator, the variation of fo as a function of IO (= IA = IB) is plotted in Fig.8. 6 Concluding Remarks A generalized scheme to realize a resistorless dual-mode sinusoidal quadrature oscillator using one CC­CTA and only two grounded capacitors is presented. The presented circuit is capable of simultaneously generating two quadrature voltage outputs and two quadrature current outputs. The frequency of oscilla­tion can be made electronically tunable by external DC biasing currents of the CCCTA. Also, the circuit sensitiv­ity study and parasitic element effects were discussed. The circuit performance is verified by PSPICE simula­tion results. 7 Acknowledgement This work was supported by the Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang (KMITL). Figure 8: Electronic tuning of fo with IO. 8 References 1. A. U. Keskin, D. Biolek, “Current-mode quadrature oscillator using current differencing transcon­ductance amplifiers (CDTA)”, IEE Proc. Circuits De­vices Syst., vol.153, no.3, pp.214-218, 2006. 2. A. Lahiri, “New current-output quadrature oscil­lator using CDTA”, IEICE Electronics Express, vol.6, no.3, pp.135-140, 2009. 3. W. Tangsrirat, W. Tanjaroen, “Current-mode sinu­soidal quadrature oscillator with independent control of oscillation frequency and condition using CDTAs”, Indian J. Pure App. Phys., vol.48, pp.363-366, 2010. 4. S. Maheshwariang, B. Chaturvedi, “High output impedance CMQOs using DVCCs and ground­ed components”, Int. J. Circ. Theor. Appl., vol.39, pp.427-435, 2011. 5. N. Minhaj, “Current conveyor-based voltage-mode two-phase and four-phase quadrature os­cillator”, Int. J. Electron., vol.94, no.6-8, pp.663-669, 2007. 6. W. Tangsrirat, S. Pisitchalermpong, “CDBA-based quadrature sinusoidal oscillator”, Frequenz, vol.61, no.3-4, pp.102-104, 2007. 7. W. Tangsrirat, D. Prasertsom, T. Piyatat, W. Surakampontorn, “Single-resistance-controlled quadrature oscillator using current differencing buffered amplifiers”, Int. J. Electron., vol.95, no.11, pp.1119-1126, 2008. 8. W. Tangsrirat, T. Pukkalanun, W. Surakampon­torn, “CDBA-based universal biquad filter and quadrature oscillator”, Active Passive Electron. Comp., vol.2008, Article ID 24717, 6 pages, doi : 10.1155/2008/247171, 2008. 9. S. Maheshwariang, J. Mohan, D. S. Chauhan, “High input impedance voltage-mode universal filter and quadrature oscillator”, J. Circuits Syst. Com­put., vol.19, no.7, pp.1597-1607, 2010. 10. N. Herencsar, K. Vrba, J. Koton, A. Lahiri, “Realisa­tions of single-resistance-controlled quadrature oscillator using a generalized current follower transconductance amplifier and a unity-gain volt­age-follower”, Int. J. Electron., vol.97, no.8, pp.897-906, 2010. 11. S. Maheshwariang, I. A. Khan, “Novel single resist­ance controlled quadrature oscillator using two CDBAs”, J. Active Passive Electron. Devices, vol.2, pp.137-142, 2007. 12. A. Lahiri, “Novel voltage/current-output quad­rature oscillator using current differencing transconductance amplifier”, Analog Integr. Circ. Sig. Process., vol.61, no.2, pp.199-203, 2009. 13. A. Lahiri, “Explicit-current-output quadrature os­cillator using second-generation current convey­or transconductance amplifier”, Radioengineering, vol.18, no.4, pp.522-526, 2009. 14. J. W. Horng, C. L. Hou, C. M. Chang, H. P. Chou, C. T Lin, Y. H. Wen, “Quadrature oscillator with ground­ed capacitors and resistors using FDCCIIs”, ETRI Journal, vol.28, no.4, pp.486-494, 2006. 15. W. Jaikla, M. Siripruchyanun, J. Bajer, D. Biolek, “A simple current-mode quadrature oscillator us­ing single CDTA”, Radioengineering, vol.17, no.4, pp.33-40, 2008. 16. J. Jin, C. Wang, “Single CDTA-based current-mode quadrature oscillator”, Int. J. Electron. Commun. (AEU), vol.66, pp.933-936, 2012. 17. M. Siripruchyanun, W. Jaikla, “Current controlled current conveyor transconductance amplifier (CCCCTA): a building block for analog signal pro­cessing”, Electrical Engineering, vol. 90, no. 6, pp. 443-453, 2008. 18. R. Prokop, V. Musil, “New modern circuit block CCTA and some its applications”, Proceedings of the Fourteenth International Scientific and Applied Science Conference-Electronics (ET-2005), Book 5, Sofia, pp.93-98, 2005. 19. R. Sotner, J. Jerabek, R. Prokop, K. Vrba, “Current gain controlled CCTA and its application in quad­rature oscillator and direct frequency modulator”, Radioengineering, vol.20, no.1, pp.317-326, 2011. 20. W. Jaikla, A. Noppakarn, S. Lawanwisut, “New gain controllable resistor-less current-mode first order allpass filter and its application”, Radioengineer­ing, vol.21, no.1, pp.312-316, 2012. 21. W. Jaikla, S. Siripongdee, P. Suwanjan, “MISO cur­rent-mode biquad filter with independent con­trol of pole frequency and quality factor”, Radio­engineering, vol.21, no.3, pp.886-891, 2012. 22. W. Tangsrirat, “Simple BiCMOS CCCTA design and resistorless analog function realization”, The Sci­entific World Journal, vol. 2014, Article ID 423979, 7 pages, http://dx.doi.org/10.1155/2014/423979, 2014. 23. R. C. Dorf, R. H. Bishop, Modern Control Systems, 11th-edition, New Jersey, Pearson prentice Hall, 2008. 24. S. S. Gupta, R. Senani, “State variable synthesis of single resistance controlled grounded capacitor oscillator using only two CFOAs”, IEE Proc. Circuits Devices Syst., vol.145, no.2, pp.135-138, 1998. 25. W. Tangsrirat, T. Pukkalanun, “Structural genera­tion of two integrator loop filters using CDTAs and grounded capacitors”, Int. J. Circ. Theor. Appl., vol.39, no.1, pp.31-45, 2011. Arrived: 02. 07. 2016 Accepted: 01. 09. 2016 W. Tangsrirat; Informacije Midem, Vol. 46, No. 3(2016), 130 – 135 (a) (b) Figure 1: The CCCTA. (a) circuit symbol (b) equivalent circuit. Figure 2: BiCMOS realization of the CCCTA. W. Tangsrirat; Informacije Midem, Vol. 46, No. 3(2016), 130 – 135 Figure 3: Proposed dual-mode sinusoidal quadrature oscillator. W. Tangsrirat; Informacije Midem, Vol. 46, No. 3(2016), 130 – 135 (a) (b) Figure 4: Practical model of the CCCTA including para­sitic elements. Figure 5: Simulated time-doamin responses for v1 and v2. (a) initial-stage responses, (b) steady-state responses W. Tangsrirat; Informacije Midem, Vol. 46, No. 3(2016), 130 – 135 (a) (a) (b) Figure 6: Simulated time-doamin responses for i1 and i2. (a) initial-stage responses, (b) steady-state responses (b) Figure 7: Simulated frequency spectrums of the pro­posed quadrature oscillator of Fig.3. (a) for v1 and v2, (b) for i1 and i2 W. Tangsrirat; Informacije Midem, Vol. 46, No. 3(2016), 130 – 135 Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 136 – 141 Solution-derived Ba0.5Sr0.5TiO3 thin-film capacitors in metal-insulator-metal configuration Tanja Pečnik1,2, Sebastjan Glinšek3, Brigita Kmet1, Barbara Malič1,2 1Electronic Ceramics Department, Jožef Stefan Institute, Ljubljana, Slovenia 2Jožef Stefan International Postgraduate School, Ljubljana, Slovenia 3CEA Grenoble, LETI, Minatec Campus, Grenoble, France Abstract: The Ba0.5Sr0.5TiO3 (BST 50/50) thin films with the thicknesses ~250 nm were deposited on polycrystalline alumina substrates by Chemical Solution Deposition. The films were prepared by the multi-step annealing process at 750 °C, 800 °C and 900 °C and the effect of the annealing temperature on the phase composition, microstructure and dielectric properties of the films was studied. All BST 50/50 films crystallize in a pure perovskite phase after heating in a rapid thermal annealing furnace. The microstructure of the film annealed at 750 °C is granular with ~30 nm sized grains. As the annealing temperature increases to 800 °C the granular microstructure remains and the average lateral grain size increases to ~70 nm, while the film annealed at 900 °C consists of predominantly columnar grains with the average lateral size ~100 nm. The kHz-range dielectric permittivity increases from 350 for the film annealed at 750 °C to 480 for the film annealed at 900 °C. Keywords: (Ba,Sr)TiO3; thin films; microstructure; dielectric properties Tankoplastni kondenzatorji na osnovi Ba0.5Sr0.5TiO3, pripravljenega s sintezo v raztopini, s konfiguracijo kovina - dielektrik - kovina Izvleček: Tanke plasti Ba0.5Sr0.5TiO3 (BST 50/50) z debelino ~250 nm smo pripravili na podlagah polikristaliničnega aluminijevega oksida s sintezo v raztopini. Vzorce smo pripravili z večstopenjskim segrevanjem pri temperaturah 750 °C, 800 °C in 900 °C in raziskovali vpliv temperature segrevanja na fazno sestavo, mikrostrukturo in dielektrične lastnosti plasti. Ugotovili smo, da vse plasti BST 50/50 kristalizirajo v čisti perovskitni fazi in da je mikrostruktura plasti, segretih pri temperaturi 750 °C finozrnata, s ~30 nm velikimi zrni. Z višjo temperaturo segrevanja, 800 °C, se je povprečna velikost zrn povečala na ~70 nm. Mikrostrukturo plasti, žganih pri 900 °C, sestavljajo pretežno stebričasta zrna s povprečno lateralno velikostjo ~100 nm. Dielektričnost plasti, izmerjena v kHz frekvenčnem območju, se je s povišanjem temperature segrevanja s 750 °C na 900 °C povečala s 350 na 480. Ključne besede: (Ba,Sr)TiO3; tanke plasti; mikrostruktura; dielektrične lastnosti * Corresponding Author’s e-mail: tanja.pecnik@ijs.si 1 Introduction Barium strontium titanate BaxSr1-xTiO3, x=0-1 (BST) is a complex perovskite material, whose phase transition temperature (Curie temperature) from paraelectric to ferroelectric phase is tuned by the Ba/Sr ratio, from ~0 K for x=0 to ~400 K for x=1. Consequently also the dielectric properties of BST are tuned by the compo­sition. In the paraelectric phase, yet close above the Curie temperature, the BST exhibits high dielectric per­mittivity and tunability, i.e. electric-field dependence of dielectric permittivity, but also low dielectric losses in GHz frequency range, which makes it suitable for the use in tunable microwave devices [1], [2]. In the case of solution-derived thin films different fac­tors such as film thickness, grain size and shape [3–7], porosity [8], residual stress [6], [9], interaction with the electrodes [1], etc., strongly modify the response of the films and therefore their effect should be considered. For example Sinnamon et al. [10] prepared BST 50/50 thin films with the thicknesses in the range from 15 nm to 1.5 µm by pulsed laser deposition on SrRuO3/MgO substrates. The authors showed that as the film thick­ness increased from 15 nm to 1.5 µm the respective av­erage lateral grain size increased from 80 nm to 460 nm. Consequently, the dielectric permittivity, measured at 10 kHz, strongly increased from around 50 for the thin­nest to 650 for the thickest film. Aygün et al. [7] studied the influence of the annealing process, i.e. one-, two- or multi-step, on the microstructure of ~550-nm-thick BaTiO3 thin films deposited by spin-coating on copper foils. When the films were prepared by one-step an­nealing, where multiple deposits were annealed only once at 900 °C, the films consisted of a granular micro­structure with ~100 nm large grains and fine pores be­tween the grains. The films prepared by the multi-step annealing, where each of many deposits was annealed separately at 900 °C, consisted of a dense and columnar microstructure with the average lateral grain size ~185 nm. The authors found that the change of the granular to the columnar microstructure and a reduced level of porosity strongly influenced the kHz-range dielectric permittivity (measured at room temperature) of the films; the dielectric permittivity increased from ~1500 for the BaTiO3 film with the granular microstructure to above 3000 for the film with the columnar microstruc­ture. In this work we focused on the preparation of Ba0.5Sr0.5TiO3 (BST 50/50) thin-film capacitors in a metal-insulator-metal configuration on platinized alumina substrates. We studied the influence of the annealing temperature on the phase composition, microstructure and dielectric properties of the films, measured in the kHz range and at room temperature. 2 Materials and methods The BST 50/50 coating solution was synthesized from the earth-alkaline acetates (Ba(CH3COO)2, 99.999 %, Alfa Aesar, Sr(CH3COO)2, 99.81 %, Alfa Aesar) and Ti-butox­ide (Ti(OC4H8)4, 99.61 %, Fluka). The acetates were dried before use and then dissolved in acetic acid (100 %, Mer­ck) and Ti-butoxide was diluted by the 2-methoxyetha­nol (CH3OCH2CH2OH, 99.3+ %, Sigma Aldrich). The two solutions were mixed for 2 hours at room temperature and the concentration of the solution was adjusted to 0.25 M. Prior deposition of the films the platinum with the thickness of ~100 nm was RF-sputtered on polished alumina substrates (99.6 %, 3.95 g/cm3, 25.4 mm x 25.4 mm x 0.26 mm, Coorstek). The BST 50/50 solution was then deposited on the substrates by spin-coating, fol­lowed by drying at 200 °C for 2 min and pyrolysis at 350 °C for 2 min. After each deposition-drying-pyrolysis step the films were heated in a rapid thermal annealing fur­nace (LPT, TM100-BT) at temperatures between 750 °C and 900 °C with the heating rate of 15 K/s. The time of annealing of the first deposit was 15 min, intermediate deposits were annealed for 5 min and the final deposit for 60 min. The deposition-drying-pyrolysis-annealing steps (multi-step annealing) were repeated seven times to reach the final thickness of ~250 nm. The phase composition of all BST 50/50 thin films was determined by PANalytical X’Pert PRO MPD X-ray dif­fractometer (XRD) with CuK.1 radiation. The XRD pat­terns were recorded in a 2. region from 10 ° to 50 ° with the step of 0.017 ° and the exposure time of 100 s. The surface and cross-section microstructures of the films were analyzed with a field-emission scanning electron microscope (FE-SEM, JSM-7600F, JEOL). The average lateral grain sizes of the BST 50/50 films were determined by the linear-intercept method based on the FE-SEM surface micrographs. For investigation of dielectric properties in the kHz fre­quency range Cr/Au top electrodes with a diameter of 0.4 mm were deposited by magnetron sputtering (5 Pa, Milano, Italy). Capacitance-voltage characteristics, measured at 100 kHz, were recorded with the following DC biasing scheme: 0 V › + 5 V › 0 V› - 5 V › 0 V. 3 Results and discussion 3.1 Phase composition The XRD patterns of BST 50/50 films annealed at 750 °C and 900 °C are shown in Figure 1. The platinized alumina substrate is added as a reference. Since the intensities of the substrate are much higher than the intensities of the perovskite BST 50/50 phase, the peaks belonging to the substrate were reduced and are denoted by *. According to the XRD analysis all films (not shown here for the film annealed at 800 °C) crystallize in a randomly oriented perovskite phase. With increasing annealing temperature the intensities of the perovskite diffrac­tion peaks increase and the full width at half maximum decreases, indicating improved crystallinity of the films and larger crystallite sizes. Figure 1: XRD patterns of the BST 50/50 films prepared on platinized alumina substrate at 750 °C and 900 °C. The peaks corresponding to the perovskite phase are denoted with the Miller indices [11]. The pattern of the substrate is also shown as a reference. * - reduced peaks of the substrate. 3.2 Microstructure The FE-SEM cross-section and plan-view micrographs of BST 50/50 films annealed at temperatures between 750 °C and 900 °C are presented in Figures 2 and 3. The thickness of the film annealed at 750 °C, determined from the cross-section micrograph, is 260 nm and de­creases to 210 nm as the annealing temperature in­creases to 900 °C. The decrease of the film thickness with increasing annealing temperature, shown also in Figure 4, indicates densification of the films. The film annealed at 750 °C consists of equiaxed grains with the average lateral size of approximately 30 nm. The surface micrograph presented in Figure 3 shows that the microstructure of the film is uniform with some fine pores between the grains. A similar granular microstructure with the grains of a few tens of nm has been commonly observed in the case of the solution-derived BaTiO3, SrTiO3 and BST thin films prepared by one- or two-step annealing processes and forms via predominantly homogenous nucleation mechanism [5], [12], [13]. With increasing the annealing temperature to 800 °C the average lateral grain size increases to approxi­mately 70 nm and the porosity and pore size decrease, as is observed from the surface microstructure of the film in Figure 3. With further increase of the anneal­ing temperature to 900 °C the average lateral grain size increases to almost 100 nm, the microstructure is uniform, dense and predominantly columnar with the grains that extend though the whole film thickness, as is shown in Figure 2. The dependence of the grain size on the annealing temperature is shown also in Figure 4; evidently the microstructure is coarsening in parallel with the enhanced densification, evidenced as the de­crease of the film thickness. The dense microstructure of the film annealed at 900 °C is related to the multi-step heat treatment where each deposit is annealed after drying and pyrolysis, which was also reported for solution-derived BaTiO3, SrTiO3 and BST thin films pre­pared by the multi-step annealing process by different research groups [5], [7], [14], [15]. 3.3 Dielectric properties Dielectric permittivity and losses of the BST 50/50 thin films annealed at temperatures between 750 °C and 900 °C are plotted in Figure 5. The dielectric permittiv­ity and losses of the BST 50/50 film annealed at 750 °C are 350 and 0.037, measured at 100 kHz and room tem­perature. With increasing the annealing temperature to 900 °C the dielectric permittivity increases to 480 and the losses remain similar, around 0.042. We connect the increase of the dielectric permittivity of the BST 50/50 films with increasing annealing temperature to the change of the granular and porous to the columnar and dense microstructure of the films, which is consist­ent with observations from the literature [5], [7], [15]. The voltage / electric-field dependence of the permit­tivity and dielectric losses of BST 50/50 film, which was annealed at 750 °C, measured at 100 kHz and 300 K, is presented in Figure 6. The tunability, expressed as the ratio of the permittivity at 0 V and 5 V, is 1.3 (23 %). Figure 5: Dielectric permittivity .` and dielectric losses tan. of BST 50/50 thin films annealed at temperatures (Tanneal) between 750 °C and 900 °C. The dielectric prop­erties were measured at 100 kHz and at room tempera­ture. Figure 6: The voltage (and field) dependence of the di­electric permittivity and losses for the BST 50/50 film annealed at 750 °C, measured at 100 kHz and room temperature. A hysteresis is observed in both curves (see Figure 6) as well as an increase of the dielectric losses as the electric field exceeds -80 kV/cm. The origin for the hysteresis in the films in the paraelectric phase could be related to the presence of polar-nano regions [16] or to the pres­ence of oxygen vacancies and space charges at the interface between the film and the substrate and/or at the grain boundaries [17]. However, explaining this phenomenon by a specific mechanism would require a further study. 4 Conclusions The effect of the annealing temperature on the phase composition, microstructure and dielectric properties of solution-derived BST 50/50 films prepared on plati­nized alumina substrates was studied. According to the XRD analysis all films crystallized in a pure perovskite phase after rapid annealing at temperatures between 750 °C and 900 °C. The FE-SEM analysis revealed that the film prepared at 750 °C was 260 nm thick and that the thickness decreased to 210 nm with increasing an­nealing temperature to 900 °C, indicating improved densification. The film annealed at 750 °C consisted of approximately 30-nm-sized equiaxed grains. The sur­face microstructure was uniform and some fine pores were observed between the grains. As the anneal­ing temperature increased to 800 °C the grains were around 70 nm in size and the porosity and pore size decreased. When the BST 50/50 film was annealed at higher temperature, i.e. 900 °C, it consisted of columnar grains with average lateral grain size around 100 nm. The dielectric permittivity of the film annealed at 750 °C was 350 and it increased to 480 with increasing annealing temperature to 900 °C, which we relate to the change of the grain size and shape and reduced level of poros­ity. 5 Acknowledgments This work was supported by the Slovenian Research Agency (P2-0105, PR-05026 and J2-5482). 6 References 1. A. K. Tagantsev, V. O. Sherman, K. F. Astafiev, J. Ven­katesh, and N. Setter, “Ferroelectric Materials for Microwave Tunable Applications,” Journal of Elec­troceramics, vol. 11, no. 1, pp. 5–66, 2004. 2. P. Bao, T. J. Jackson, X. Wang, and M. J. Lancaster, “Barium strontium titanate thin film varactors for room-temperature microwave device applica­tions,” Journal of Physics D: Applied Physics, vol. 41, no. 6, pp. 063001–1–21, 2008. 3. M. H. Frey, Z. Xu, P. Han, and D. A. Payne, “The role of interfaces on an apparent grain size effect on the dielectric properties for ferroelectric barium titanate ceramics,” Ferroelectrics, vol. 206–207, pp. 337–353, 1998. 4. K. Kageyama, A. Sakurai, A. Ando, and Y. Sakabe, “Thickness effects on microwave properties of (Ba,Sr)TiO3 films for frequency agile technologies,” Journal of the European Ceramic Society, vol. 26, no. 10–11, pp. 1873–1877, 2006. 5. S. Hoffmann and R. Waser, “Control of the mor­phology of CSD-prepared (Ba,Sr)TiO3 thin films,” Journal of the European Ceramic Society, vol. 19, no. 6–7, pp. 1339–1343, 1999. 6. T. Pečnik, S. Glinšek, B. Kmet, and B. Malič, “Com­bined effects of thickness, grain size and residual stress on the dielectric properties of Ba0.5Sr0.5TiO3 thin films,” Journal of Alloys and Compounds, vol. 646, pp. 766–772, 2015. 7. S. M. Aygu¨n, J. F. Ihlefeld, W. J. Borland, and J.-P. Maria, “Permittivity scaling in Ba1-xSrxTiO3 thin films and ceramics,” Journal of Applied Physics, vol. 109, no. 3, pp. 034108–1–5, 2011. 8. T. Ostapchuk, J. Petzelt, I. Rychetský, V. Porokhon­skyy, B. Malič, M. Kosec, and P. Vilarinho, “Influence of porosity on the dielectric response and central-mode dynamics in PbZrO3 ceramics,” Ferroelec­trics, vol. 298, no. 1, pp. 211–218, 2004. 9. E. A. Fardin, A. S. Holland, K. Ghorbani, E. K. Ak­dogan, W. K. Simon, A. Safari, and J. Y. Wang, “Polycrystalline Ba0.6Sr0.4TiO3 thin films on r-plane sapphire: Effect of film thickness on strain and di­electric properties,” Applied Physics Letters, vol. 89, no. 18, pp. 182907–1–3, 2006. 10. L. J. Sinnamon, M. M. Saad, R. M. Bowman, and J. M. Gregg, “Exploring grain size as a cause for ‘dead-layer’ effects in thin film capacitors,” Applied Physics Letters, vol. 81, no. 4, pp. 703–705, 2002. 11. “PDF 00-039-1395.” JCPDS-International Center for Diffraction Data, Newton Square, 2002. 12. R. W. Schwartz, P. G. Clem, J. A. Voigt, E. R. Byhoff, M. Van Stry, T. J. Headley, and N. A. Missert, “Con­trol of microstructure and orientation in solution-deposited BaTiO3 and SrTiO3 thin films,” Journal of the American Ceramic Society, vol. 82, no. 9, pp. 2359–2367, 1999. 13. B. Malič, I. Boerasu, M. Mandeljc, M. Kosec, V. Sher­man, T. Yamada, N. Setter, and M. Vukadinovic, “Processing and dielectric characterization of Ba0.3Sr0.7TiO3 thin films on alumina substrates,” Journal of the European Ceramic Society, vol. 27, no. 8–9, pp. 2945–2948, 2007. 14. C. Jia and K. Urban, “Microstructure of columnar-grained SrTiO3 and BaTiO3 thin films prepared by chemical solution deposition,” Journal of Materials Research, vol. 13, no. 8, pp. 2206–2217, 1998. 15. K. Kageyama, T. Hosokura, T. Nakaiso, and H. Takagi, “Dielectric properties of (Ba,Sr)TiO3 thin films with varied microstructures prepared by the chemical solution deposition method for thin-film capacitors and ferroelectric varactors.,” IEEE transactions on ultrasonics, ferroelectrics, and fre­quency control, vol. 57, no. 10, pp. 2198–204, 2010. 16. H. W. Jang, A. Kumar, S. Denev, M. D. Biegalski, P. Maksymovych, C. W. Bark, C. T. Nelson, C. M. Folk­man, S. H. Baek, N. Balke, C. M. Brooks, D. A. Tenne, D. G. Schlom, L. Q. Chen, X. Q. Pan, S. V. Kalinin, V. Gopalan, and C. B. Eom, “Ferroelectricity in strain-free SrTiO3 thin films,” Physical Review Letters, vol. 104, no. 19, pp. 197601–1–4, 2010. 17. X. H. Zhu, L. P. Yong, H. F. Tian, W. Peng, J. Q. Li, and D. N. Zheng, “The origin of the weak ferroelectric-like hysteresis effect in paraelectric Ba0.5Sr0.5TiO3 thin films grown epitaxially on LaAlO3,” Journal of Physics: Condensed Matter, vol. 18, no. 19, pp. 4709–4718, 2006. Arrived: 31. 08. 2016 Accepted: 22. 09. 2016 T. Pečnik et al; Informacije Midem, Vol. 46, No. 3(2016), 136 – 141 T. Pečnik et al; Informacije Midem, Vol. 46, No. 3(2016), 136 – 141 Figure 2: The FE-SEM cross-section micrographs of the BST 50/50 films annealed at temperatures between 750 °C and 900 °C. T. Pečnik et al; Informacije Midem, Vol. 46, No. 3(2016), 136 – 141 Figure 3: The FE-SEM surface micrographs of the BST 50/50 films annealed at temperatures between 750 °C and 900 °C. Figure 4: The dependence of the film thickness t and grain size GS on the annealing temperature Tanneal of the BST 50/50 thin films. T. Pečnik et al; Informacije Midem, Vol. 46, No. 3(2016), 136 – 141 T. Pečnik et al; Informacije Midem, Vol. 46, No. 3(2016), 136 – 141 Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 142 – 148 Temperature and Injection Current dependent Electroluminescence for Evaluation of Single-Junction Single-Segment GaAs Laser Power Converter Rok Kimovec1, Henning Helmers2, Andreas W. Bett2 and Marko Topič1 1University of Ljubljana, Faculty of Electrical Engineering, Ljubljana, Slovenia 2Fraunhofer Institute for Solar Energy Systems ISE, Freiburg, Germany Abstract: The spatial electroluminescence intensity and spectral measurements of photovoltaic devices under forward bias have proved to be fast and reliable characterization tools. They enable quick evaluation of material and manufacturing quality and provide information linked to local performance of photovoltaic devices in different operating conditions. In this work, EL images of single-junction single-segment GaAs laser power converters (LPC) and their emission spectra depending on the injection current and LPC temperature are presented and analyzed. A shift of the EL emission peak to smaller energies and a decrease in EL intensity with increasing temperature are observed in spectrally resolved EL measurements. Negative temperature coefficients dEL/dT of EL intensity depending on the injection current are extracted from spatially resolved EL measurements. EL images of the LPCs under low and high injection currents are presented and evaluated. Keywords: electroluminescence; laser power converter; Power-over-Fiber; power-by-light; spatial electroluminescence; spectral electroluminescence; LPC Vpliv temperature in vzbujalnega toka na elektroluminiscenco enosegmentnih enospojnih pretvornikov laserske moči Izvleček: Meritve prostorske in spektralne elektroluminiscence fotovoltaičnih struktur so se izkazale za zanesljivo in hitro karakterizacijsko orodje. Omogočajo nam vpogled v kvaliteto uporabljenega materiala in izdelave struktur ter nam podajo možnost vpogleda v spreminjanje kazalcev učinkovitosti pri spreminjajočih delovnih pogojih. V tem delu so predstavljene meritve in analiza prostorske in spektralne elektroluminiscence v odvisnosti od delovne temperature in vzbujalnega toka za primer enosegmentnega enospojnega pretvornika laserske moči izdelanega iz GaAs. Z višanjem temperature se intenziteta izsevane svetlobe zmanjša, izsevani spekter pa se premakne proti nižjim energijam. Iz analize prostorske elektroluminiscence je izračunan negativni temperaturni koeficient intenzitete izsevane svetlobe dEL/dT pri različnih vzbujalnih tokovih. Predstavljena je analiza slik prostorske elektroluminiscence za primer majhnega in velikega vzbujalnega toka. Ključne besede: elektroluminiscenca; pretvornik laserske moči; Moč-po-vlaknu; prostorska elektroluminiscenca; spektralna elektroluminiscenca * Corresponding Author’s e-mail: rok.kimovec@fe.uni-lj.si 1 Introduction Electroluminescence (EL) in photovoltaic (PV) devices is a phenomenon occurring due to radiative recombi­nation of electrically excited charge carriers, resulting in the emission of photons from the PV device struc­ture. The measurement of EL, coupled with theoretical knowledge, has become an important tool in assessing performance of PV devices in both scientific and indus­trial environments [1–4], due to fast acquisition times of modern imaging devices. For purposes of Power-over-Fiber (PoF) ( or “power-by-light”) [5, 6], where optical energy in the form of laser light is transferred through electrically nonconductive optical fiber, special types of PV devices (laser power converters (LPC)) are utilized to convert monochromat­ic light to electricity [7, 8]. As for all other PV devices, performance of LPCs is upwards limited [9] and it can be evaluated to some extent by the use of spatially and spectrally resolved EL measurements. The aim of this work is to gain insight into spatially and spectrally resolved characterization techniques and performance of LPCs under various operation temper­atures and current densities. 2 Motivation PoF is an emerging technology used for powering vari­ous electronic devices in extreme environments, where benefits such as galvanic isolation and electromag­netic compatibility overcome the additional cost and complexity of such power supply systems [10, 11]. To gain in-depth understanding of PoF systems, evalu­ation of individual components has to be undertaken. In this work we concentrate on assessing the per­formance of LPCs using advanced spatially resolved characterization techniques. The LPC is the part of the system which is converting monochromatic light into electricity, thus providing electrical energy to an elec­tronic device at a remote location. Since such systems are used for powering a variety of electronic devices (e.g. sensors [12], cameras [13], actuators, etc.) in many different environments (e.g. space [14], high magnetic fields [15], etc.), they must perform well in different operating conditions, such as variation in temperature and power requirements for different electronic de­vices. In order to understand the behavior and performance of LPCs in different working conditions better, an analy­sis of EL at various temperatures and injection currents has been performed and is presented in this paper. 3 Description of specimen The investigated LPC sample is a single-junction single-segment (SJ-SS) GaAs photovoltaic device fabricated at Fraunhofer ISE. The active region is a 3.65 µm thick pin-stack with an n-type emitter of 120 nm thickness. It employs 25 nm AlInP window-layer on top of the emitter which acts as a passivation layer (front surface field). The p-doped base layer follows a back surface field (BSF) layer in order to passivate the rear side of the active solar cell. A schematic representation of the de­vice structure is shown in Figure 1 (a). The chip area is 3 mm x 3 mm square with front contact circular busbar design, which covers 2 mm2 of the area. An image of the device under test (DUT) is shown in Figure 1 (b). (a) (b) Figure 1: (a) Schematic of the investigated LPC. (b) Top view of 3 mm x 3 mm square LPC sample. Even though the device is wire bonded to a TO-header in two corners only the top right connection was used for measurement. 4 Experimental setup In the following the experimental setup and proce­dures are described, that have been used to measure the spatially and spectrally resolved EL of SJ–SS GaAs LPC at the Laboratory of Photovoltaics and Optoelec­tronics at the Faculty of Electrical Engineering, Univer­sity of Ljubljana. The results of the measurements of a SJ–SS GaAs LPC are presented and discussed. 4.1 General measurement description To perform EL measurements, the LPC sample has been mounted on a temperature controlled chuck and placed into a light proof box, ensuring that only light irradiated by the LPC is measured. Measurements of injection current and voltage were realized by four-wire connection at various injection current values, provided by an HP E363X series labora­tory power supply operating in constant current mode. Current and voltage signals were measured with a pair of HP 34401A multimeters. The measurement procedure for both spatial and spec­tral EL was as follows: - Setting of desired injection current - Setting of desired temperature - Setting of acquisition time once the temperature was stable, to avoid saturation of the camera/spectrometer - Measurement of spatial/spectral EL 4.2 Spatially resolved measurement setup To obtain spatially resolved images a FLI MLx285 scien­tific camera was used, employing a Sony ICX285 CCD sensor (actively cooled to 0°C). The primary lens on the camera was a Schneider Optics NR56-534. Since the magnification with the primary lens was insufficient to take images of the small area LPC, a microscope objec­tive was mounted on the primary lens with use of tube extenders and a custom made adapter. The described setup enables to acquire sharp EL images of small area LPCs with a resolution higher than 500x500 pixels. 4.3 Spectrally resolved measurement setup EL spectra were measured with an Ocean Optics HR4000 spectrometer equipped with 600 µm core mul­timode optical fiber and cosine corrector. The setup was calibrated with a NIST traceable Ocean Optics LS-1-CAL light source prior to the EL measurements. The optical fiber input connector equipped with the cosine corrector was placed in close proximity above the LPC. At large injection currents the current distribution is ex­pected to be non-uniform, leading to large differences of EL irradiation through the front surface of the LPC. Care was taken to assure that all irradiated light from the device under test was coupled into the fiber. Even though the spectrum acquisition system was cali­brated prior to measurement, it should be noted that such measurement systems are insufficient to measure absolute irradiance values, so that all spectral measurements results are expressed in arbitrary units. Still, they can serve for rela­tive comparison of LPCs at varying operating conditions. 5 Results 5.1 Spatially resolved measurements To perform analysis of images captured by the CCD camera, two corrections of the raw data were per­formed. First, the dark image was subtracted from all images. Second, individual pixel values were divided by acquisition time, resulting in each pixel value pre­sented in counts per second acquired by the CCD. Fur­ther details on the measurement procedure and image manipulation procedure can be found elsewhere [16]. Acquired images from spatially resolved EL enable to perform two different types of analysis. First it is a quali­tative analysis in which the material homogeneity can be evaluated at low injection currents. At high injection currents on the contrary, the influence of the busbar and sheet series resistances on the current distribu­tion can be seen through the top layers of the device. The results of such an analysis are shown in Figure 2 and Figure 3 for two different injection currents (1 mA, 400 mA). Under the injection current of 1 mA (Figure 2) the LPC shows a mostly uniform electroluminescence across the whole cell area, which can be attributed to a homogeneous material quality of the GaAs crystal. With an injection current of 400 mA (Figure 3), the measured EL intensity distribution across the LPC area decreases from the top right corner to the bottom left corner. This behavior can be explained by non-ideal contacting of the specimen. Opposed to an ideal con­tacting in each corner of the device, wire bonds were only placed in the top right and bottom right corner (compare Fig. 1b), of which only the three wire bonds in the top right corner were electrically contacted dur­ing the measurements described in this work. Thus, the injected current needs to redistribute from the top right corner to the rest of the busbar metallization. For high current conditions the narrowest sections of the busbar represent a significant bottleneck to the current flow, resulting in a significant voltage drop at those points. For a device contacted in all four corners, this voltage drop would not occur and the EL profile is expected to be radially symmetrical, with the EL signal decaying from the border to the center of the active area. A second effect is seen as a decay of EL from the top right corner to the center of the LPC. This effect can be explained by a voltage drop at the grid and top surface resistance and would stay the same, regardless the connection of the LPC to the external contacts. A combination of both effects, can be clearly seen in nor­malized EL along the line scan, marked with a cyan line in Figure 3. To mitigate resistance losses from lateral conduction above the pn-junction, recent designs of LPCs employ a so-called lateral conduction layer [17]. This layer is composed of a highly doped material with a bandgap larger than the photon energy of the im­pinging light – making it transparent for the used laser wavelength. In addition to the qualitative assessment of the current distribution, the spatially resolved EL measurements can also be assessed for a relative quantitative analy­sis. Therefore, the mean pixel value of the circular ac­tive region (marked with a green circle in Figure 3) is calculated and used for normalization of all EL images (various injection currents and temperatures). In Figure 4 the resulting counts per second are plotted, as re­corded by the CCD camera after subtraction of the dark image. Consequently, this value corresponds to the de­pendency of the EL intensity on the injection current and temperature. The shapes of the EL curves can be described by a pow­er-law current dependence [18]. (1) kCCD is the efficiency of CCD sensor and optics and C(T) and b(T) are in general temperature dependent coeffi­cients. Similar behavior of EL vs. current was previously observed for various photovoltaic devices [16], [19, 20]. Figure 4: Plot of EL intensity vs. injection current from SJ-SS GaAs LPC extracted from mean pixel values of processed EL images at various temperatures. In Figure 6 the EL intensity is plotted normalized to the value at T=10°C as a function of temperature. As can be seen, for all injection currents the EL intensity de­creases with increasing temperature. The correspond­ing negative temperature coefficient depends on in­jection currents. Increasing temperature has a larger impact on EL for lower injection currents as seen in Figure 5. Temperature coefficients dEL/dT change from -1.5 %/°C for injection current of 1 mA to -0.5 %/°C for injection current of 400 mA. Figure 5: Temperature dependence of the EL intensity extracted from mean pixel values of processed EL im­ages at different injection currents. 5.2 Spectrally resolved measurements While spatially resolved EL characterization enables quick, qualitative analysis, it lacks information on the spectral distribution of the emitted light. Since the spectrum of emitted light corresponds to the bandgap of the direct semiconductor, it provides information about the change of the absorber material’s bandgap with temperature. As can be seen in Figure 6, increas­ing temperature results in two different effects: Firstly, the EL emission peak is shifted to lower energies, as expected due to the dilation of the crystal lattice and corresponding decrease in bandgap described by the following equation [21]: (2) where Eg is the bandgap and E0, . and ß are material dependent parameters. The measured temperature coefficient dEg_meas/dT = -0.46 meV/K agrees well with the published value for that temperature range dEg_pub/dT = 0.45 meV/K [22, 23]. Secondly, it provides insight into how EL emission de­creases with rising temperature. While the slope at low­er energies (i.e. below the bandgap) remains constant, the slope at higher energies (i.e. above the bandgap) decreases with increasing temperature. This has been observed and explained before [4]. In a PoF system, the LPC is illuminated by a laser diode. Since the laser diode will typically be temperature con­trolled, the wavelength and thus photon energy of the laser will be constant. With maximal absorption current generation results of the impinged monochromatic light is maximized. Figure 6: Spectral EL emission of SJ-SS GaAs LPC under constant injection current of 400 mA for various tem­peratures. Figure 7: Spectral EL emission of a SJ-SS GaAs LPC at constant temperature of 25 °C at various injection cur­rents. The red line marks the emission peak at 1.422eV (.872 nm). To ensure maximal absorption the the photon energy has to be slightly smaller than the bandgap of the LPC absorber material [7]. However, the difference should be small, as the difference in photon and bandgap en­ergy represents an energy loss. It is mainly converted Video Camera Link,” IEEE Photonics Technol. Lett., vol. 20, no. 1, pp. 39–41, Jan. 2008. 14. R. Pena and C. Algora, “One-watt fiber-based power-by-light system for satellite applications,” Prog. Photovolt. Res. Appl., vol. 20, no. 1, pp. 117–123, Jan. 2012. 15. J. G. Werthen, M. J. Cohen, T. C. Wu, and S. Widjaja, “Electrically isolated power delivery for MRI appli­cations,” in Proc. Int. Soc. Magn. Reson. Med. Annu. Meeting, Seattle, WA, 2006, p. 1353. 16. M. Bokalič, J. Raguse, J. R. Sites, and M. Topič, “Analysis of electroluminescence images in small-area circular CdTe solar cells,” J. Appl. Phys., vol. 114, no. 12, p. 123102, Sep. 2013. 17. E. Oliva, F. Dimroth, and A. W. Bett, “GaAs convert­ers for high power densities of laser illumination,” Prog. Photovolt. Res. Appl., vol. 16, no. 4, pp. 289–295, Jun. 2008. 18. K. Wang, D. Han, and M. Silver, “The Power Law Dependence of Electroluminescence Intensity on Forward Current in a-Si:H p-i-n Devices,” in Sym­posium A – Amorphous Silicon Technology - 1994, 1994, vol. 336, p. 861 (6 pages). 19. K. J. Price, A. Vasko, L. Gorrelland, and A. D. Com­paan, “Temperature-Dependent Electrolumines­cence from CdTe/CdS Solar Cells,” in Symposium B – Compound Semiconductor Photovoltaics, 2003, vol. 763, p. B5.9 (6 pages). 20. M. Seeland, R. Rösch, and H. Hoppe, “Quantita­tive analysis of electroluminescence images from polymer solar cells,” J. Appl. Phys., vol. 111, no. 2, p. 024505, Jan. 2012. 21. Y. P. Varshni, “Temperature dependence of the en­ergy gap in semiconductors,” Physica, vol. 34, no. 1, pp. 149–154, Jan. 1967. 22. C. D. Thurmond, “The Standard Thermodynamic Functions for the Formation of Electrons and Holes in Ge, Si, GaAs , and GaP,” J. Electrochem. Soc., vol. 122, no. 8, pp. 1133–1141, Aug. 1975. 23. P. Lautenschlager, M. Garriga, S. Logothetidis, and M. Cardona, “Interband critical points of GaAs and their temperature dependence,” Phys. Rev. B, vol. 35, no. 17, pp. 9174–9189, Jun. 1987. Arrived: 31. 08. 2016 Accepted: 22. 09. 2016 R. Kimovec et al; Informacije Midem, Vol. 46, No. 3(2016), 142 – 148 R. Kimovec et al; Informacije Midem, Vol. 46, No. 3(2016), 142 – 148 Figure 2: Top: Normalized EL image of a SJ-SS GaAs LPC at an injection current of 1 mA. The metallized front grid is visible as dark lines in the EL image. Bottom: Plot of the normalized EL along line scan, marked with cyan, showing an even distribution of the current. Note that the line used for the line scan is actually a rectangle, which covers a narrow area of the LPC beside the cen­tral finger. R. Kimovec et al; Informacije Midem, Vol. 46, No. 3(2016), 142 – 148 Figure 3: Top: Normalized EL image of a SJ-SS GaAs LPC with injection current 400 mA. The metallized front grid is visible as dark lines in the EL image. Bottom: Plot of normalized EL along a line scan, marked with cyan, showing a combination of various resistances on the EL profile through the active area. Note that the line used for the line scan is actually a rectangle, which covers an area of the LPC beside the central finger. The green circle marks the central active region, taken for calcula­tions of the relative quantitative analysis. R. Kimovec et al; Informacije Midem, Vol. 46, No. 3(2016), 142 – 148 R. Kimovec et al; Informacije Midem, Vol. 46, No. 3(2016), 142 – 148 Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 149 – 153 Linear Incremental Displacement Measurement System with Microtransformers Matija Podhraški1, Janez Trontelj2 1Letrika Lab d.o.o., Šempeter pri Gorici, Slovenia 2Laboratory for Microelectronics, Faculty of Electrical Engineering, Ljubljana, Slovenia Abstract: The paper discusses an inductive microsensor system for displacement measurement comprising microtransformers. The primary windings of the microtransformers are excited with an AC source with a frequency of several MHz. The microtransformers are fabricated in internal metal layers of an integrated circuit using a conventional 350 nm commercial CMOS process, along with corresponding circuits for the processing of the microtransformers’ output signals. The major advantage of such system is its cost-effectiveness due to its straightforward fabrication and the absence of the need for an external field generator, such as permanent magnets at Hall Effect encoders or a light source at optical encoders. In a linear incremental encoder application, microtransformer output signals are modulated by a metal measurement scale positioned over the integrated microsystem, resulting in a combination of amplitude and phase modulation. The integrated circuit employs a fully-differential measurement channel with three-stage amplification and a mixer implemented with a Gilbert cell: the signal is demodulated using synchronous demodulation. A prototype microsystem was designed, fabricated and evaluated, demonstrating a sensitivity of 0.99 V/mm with a copper target at an approximate microsystem-target distance of 200-250 µm. Keywords: inductive sensor; eddy-current sensor; displacement sensor; ASIC; microtransformer; linear encoder Sistem z mikrotransformatorji za inkrementalno merjenje linearnega pomika Izvleček: Prispevek obravnava induktivni mikrosenzorski sistem za merjenje pomika na osnovi mikrotransformatorjev. Primarna navitja mikrotransfomatorjev so vzbujana z izmeničnim virom frekvence nekaj MHz. Mikrotransformatorji so izdelani v internih metalnih slojih integriranega vezja, proizvedenega s konvencionalnim 350 nm komercialnim CMOS procesom, pridružena pa so jim tudi ustrezna vezja za procesiranje izhodnih signalov mikrotransformatorja. Glavna prednost takšnega sistema je njegova cenovna učinkovitost zaradi preproste izdelave in odsotnosti potrebe po zunanjem generatorju polja, kot so npr. trajni magneti pri Hallovih enkoderjih oziroma svetlobni viri pri optičnih. V aplikaciji linearnega inkrementalnega enkoderja so izhodni signali mikrotransfomatorja modulirani s kovinsko merilno letvijo, nameščeno nad integriran mikrosistem, kar se odraža v kombinaciji amplitudne in fazne modulacije. Integrirano vezje vsebuje popolno diferencialni merilni kanal s trostopenjskim ojačenjem in mešalnik, izveden z Gilbertovo celico: signal je sinhronsko demoduliran. Zasnovan, izdelan in izmerjen je bil prototipni mikrosistem z doseženo odzivnostjo 0,99 V/mm pri bakreni tarči in oddaljenosti med tarčo in senzorjem približno 200-250 µm. Ključne besede: induktivni senzorji; senzorji na vrtinčne tokove; senzorji pomika; namensko integrirano vezje; mikrotransformatorji; linearni enkoder * Corresponding Author’s e-mail: matija.podhraski@si.mahle.com M. Podhraški et al; Informacije Midem, Vol. 46, No. 3(2016), 149 – 153 1 Introduction The main difference of inductive position sensing con­cept in comparison to conventional magnetic encoders (which are based on Hall or magnetoresistive sensors) is in the use of an alternating magnetic field instead of a stationary magnetic field; sensors employ the prin­ciple of electromagnetic induction. Two major types of inductive sensors are used [1], [2]. The first type is a dual-coil structure, similar to a trans­former. The first coil is connected to an AC source, in­ducing the voltage in the second coil. If a conductive object is moved close to the coils, eddy currents are in­duced in the object. Due to the loss of energy through this mechanism, the voltage in the secondary coil is reduced [3]. The effect on the secondary voltage is ad­versary in the presence of a ferromagnetic object, im­proving the magnetic coupling between the coils [3]. The second type is based on the change of the coil in­ductance under the effect of a nearby object: if a coil is wired into a resonant circuit, its oscillation frequency changes when the object moves [2]. Inductive sensors benefit from their insensitivity to dust, which stands out as a strong advantage in an in­dustrial environment in comparison to the optical sen­sors [4]. Magnetic and optical position encoders can be fabri­cated as application-specific integrated circuits (ASICs). However, for their use, external placement of magnetic field source or light source is needed. Inductive sen­sors are free from this requirement, since they generate the high frequency magnetic field by an integrated in­ductor. In this paper, we present a microelectronic im­plementation of a prototype inductive linear position encoder, operating with a passive measurement scale. The sensor elements are realized as microtransformers with the accompanying electronics fabricated together with the microtransformers in an ASIC using an unmod­ified 350 nm CMOS process. 2 Design The discussed system operates similarly as a linear vari­able differential transformer (LVDT), as well as an eddy current sensor [1–3], [5]. The sensor is scaled to the size of a typical integrated circuit (several square millime­ters). The design of the microtransformer setup used in the sensor is shown in Figure 1. Figure 2 displays the differential operation of the microtransformer. When a full half-period of a ferromagnetic scale is positioned over the first microtransformer, the coupling between the primary and the secondary winding is the stron­gest for this microtransformer. Contrarily, the coupling is then the weakest for the second microtransformer as the void half-period is positioned over it [2], [3]. Figure 1: The structure of a microtransformer pair (P – primary, S – secondary winding) [2]. Figure 2: The differential operation of a microtrans­former pair [2]. The differential voltage of the microtransformer pair Vdiff is obtained by subtracting the secondary voltages of microtransformers Va and Vb [3]. In the described sit­uation (Figure 2), Vdiff amplitude is maximal. As the scale moves, the outputs change periodically. It should be noted that for a conductive (non-ferromagnetic) scale, the operation is adversary [5]. When a microtransform­er is completely covered with a part of non-ferromag­netic metal, its induced voltage is minimal due to en­ergy dissipation in the scale through the mechanism of eddy currents [3]. Using the presented differential principle, the signals which are common to both microtransformers in a pair (such as EMI and the capacitively transferred voltage) are subtracted [5]. The general design of the microsystem is presented in Figure 5 (a). It consists of a silicon die comprising the mi­crotransformers along with analog front-end electron­ics for the generation of the differential signal [3]. The microtransformers are fabricated using standard CMOS technology metal layers. The total layer count is four. The external dimensions of the microtransformer primary and secondary windings are 755 by 500 µm and 576 by 314 µm, respectively [3]. Therefore the scale period P is 1 mm. Each winding of a microtransformer has 45 turns: three layers with 15 turns per layer are used, while the top metal layer is used for routing the connections [3]. The winding structure for a single winding is shown in Figure 4. Such structure is used for reducing the inter-winding capacitance [3]. A model circuit of a microtrans­former is shown in Figure 3, with the accompanying component values given in Table 1. Such circuit is insuf­ficient to model the effects of the measurement scale on the output voltage of a microtransformer. So, finite element modeling was used to acquire the modulation characteristics as described in [3], [6]. Table 1: Component values in the model circuit [3]. Components Value R1 , R2 2657 . R3 , R4 1816 . L1 , L2 1.16 µH L3 , L4 658 nH C1 3.55 pF C2 3.4 fF C3 2.39 pF k1 , k2 0.429 To improve the signal-to-noise ratio of the system, the output signals of the coils with same position relative to the scale period can be summed, as shown in Figure 5 (b). The primary windings are wired in parallel [2]. Figure 5: (a) A block representation of the presented microsystem with a metal scale of period P and quad­rature output signals. (b) The implemented summation scheme [2]. The device comprises two channels shifted for a quar­ter of the scale period, i.e. quadrature output signals [3]. The quadrature principle is commonly employed in position encoders (e.g. optical [7] and Hall devices [8]), relying on (multiples of) two sensor elements with their position shifted by a half of the primary coil width (i.e. 1 of the scale period P). Observing the phase shift of the quadrature signals allows the determination of the movement direction. If the signals have a sinusoi­dal shape, the arctangent function of their amplitude ratio enables a straightforward calculation of the dis­placement inside a single half-period [3]. (1) A block diagram of a single measurement channel as implemented in the integrated circuit is shown in Fi­gure 8. A fully differential channel setup is used, with the subtraction of the positive and negative micro­transformer output signal performed at the end of the chain (Stage 3). Figure 6: The Gilbert cell mixer implemented in the ASIC [6]. The first amplifier is wideband (72 MHz GBW), employ­ing telescopic topology [3]. Then, the signal is mixed down to DC using a differential Gilbert cell CMOS mixer [6], shown in Figure 6. In the next two stages, signals are amplified at the baseband, also filtering out the re­maining HF signal components [3]. 3 Evaluation To evaluate the performance of the microsystem, it was placed on a mechanical micromanipulator controlled by a computer, which was used to displace a measure­ment scale. Two scales (Figure 7) were used: scale (1) was made by laser cutting from transformer steel sheet (0.35 mm thickness), and the second (2) was fabricated as a PCB (35 µm copper thickness) [3]. Due to the pres­ence of gel coating needed for the IC protection, the thickness between the scale and the surface of the IC was no less than 250-300 µm [3]. Figure 7: Scales used for the evaluation [3]. First, the excitation frequency and the phase of the mixing signal were swept to determine the optimal pa­rameters. The maximal peak-to-peak amplitude of the output signal was chosen as the figure of merit [3]. Figure 8: a block diagram of a single measurement channel implemented in the ASIC [3]. The output characteristics were recorded at the opti­mal excitation frequency fexc and mixing signal phase .mix for the copper and steel scale with 20 µm position­ing step. The results are given in Figure 9. The sensitiv­ity S of the microsystem is defined (Equation 2) as the change of the output peak-to-peak voltage over a scale period P [3]: (2) Sensitivities for the two scales as well as maximum and RMS values of the linearity error E are given in Table 2. Table 2: Summarized measurement results [3]. Copper scale Steel scale S (Ch. 1) 0.99 0.57 S (Ch. 2) 0.71 0.44 max (E) 18.79 33.05 rms (E) 6.89 11.32 4 Conclusion The design and the evaluation of an integrated micro­transformer linear position measurement system were demonstrated. The system was evaluated with two scale types. It was discovered that various scales have different optimal excitation frequencies and phases of the mixing signal [3]. Therefore, a system should be adaptable to support the variation of these param­eters. Considering the microtransformer sensitivity as well as the linearity error, better results were observed with the copper scale. In our future work, we intend to redesign the measure­ment channel to reduce measurement noise by moving the major part of the amplification to the first amplify­ing stage, and to implement an on-chip frequency and phase-tunable oscillator, resulting in a true single-chip linear position encoder, having a significant potential for the encoder industry due to its cost-efficiency. 5 References 1. M. Podhraški and J. Trontelj, “Design and evalu­ation of a microcoil proximity sensing microsys­tem,” Conf. 2015 Proc. 51th Int. Conf. Microelectron. Devices Mater. Workshop Terahertz Microw. Syst. Sept. 23 - Sept. 25 2015 Bled Slov., vol. 2015, pp. 95–99, 2015. 2. M. Podhraški and J. Trontelj, “An integrated micro­transformer system for displacement measure­ment,” Inf. MIDEM, vol. 46, no. 1, pp. 29–35, 2016. 3. M. Podhraški and J. Trontelj, “A Differential Mo­nolithically Integrated Inductive Linear Displace­ment Measurement Microsystem,” Sensors, vol. 16, no. 3, p. 384, Mar. 2016. 4. A. J. Fleming, “A review of nanometer resolution position sensors: Operation and performance,” Sens. Actuators Phys., vol. 190, pp. 106–126, Feb. 2013. 5. J. W. Bergqvist, Y. de Coulon, and H. de Lambilly, “Device for detecting position and movement by using magnetic field variation,” US6043644 A, 28-Mar-2000. 6. M. Podhraški, “Integrirani mikrosenzorski sistemi z mikrotuljavicami,” Univerza v Ljubljani, Ljublja­na, 2016. 7. J. Rozman and A. Pletersek, “Linear Optical Enco­der System With Sinusoidal Signal Distortion Be­low 60 dB,” IEEE Trans. Instrum. Meas., vol. 59, no. 6, pp. 1544–1549, Jun. 2010. 8. H. V. Hoang and J. W. Jeon, “Signal compensation and extraction of high resolution position for sinusoidal magnetic encoders,” in International Conference on Control, Automation and Systems, 2007. ICCAS ’07, 2007, pp. 1368–1373. Arrived: 31. 08. 2016 Accepted: 22. 09. 2016 Figure 3: A model circuit of a microtransformer [3]. M. Podhraški et al; Informacije Midem, Vol. 46, No. 3(2016), 149 – 153 Figure 4: The microtransformer winding design [3]. M. Podhraški et al; Informacije Midem, Vol. 46, No. 3(2016), 149 – 153 Figure 9: ASIC characterization results for both scale types. Results are compared to an ideal arctangent function. M. Podhraški et al; Informacije Midem, Vol. 46, No. 3(2016), 149 – 153 Journal of Microelectronics, Electronic Components and Materials Vol. 46, No. 3(2016), 154 – 159 Control of electrical conductivity in 0.7BiFeO3-0.3SrTiO3 ferroelectric ceramics via thermal treatment in nitrogen atmosphere and Mn doping Maja Makarovič1,2, Julian Walker3, Evgeniya Khomyakova1,2, Andreja Benčan1,2, Barbara Malič1,2 and Tadej Rojac1,2 1Jožef Stefan Institute, Ljubljana, Slovenia 2Jožef Stefan International Postgraduate School, Ljubljana, Slovenia 3Materials Research Institute, Pennsylvania State University, USA Abstract: In this work, the solid solution between polar BiFeO3 (BFO) and non-polar SrTiO3 (ST) with the composition 0.7BFO-0.3ST has been prepared by mechanochemical activation-assisted synthesis with particular emphasis on the characterization and control of the electrical conductivity of the resulting ceramics. According to X-ray diffraction analysis and scanning electron microscopy the incorporation of ST into BFO minimizes the formation of secondary phases, typically formed during the synthesis of unmodified BFO. The as-sintered ceramics exhibited a high electrical conductivity, which was suppressed by post-annealing in N2 atmosphere. However, this approach showed two major drawbacks: i) re-oxidation of samples and thus increase in their conductivity when annealed in air to elevated temperatures (up to ~450 °C) and ii) increased conductivity by application of high electrical fields, resulting in a strong leakage-current contribution to the measured polarization-electric-field hysteresis loops. For these reasons, in order to reduce the conductivity, we propose here an alternative approach, i.e., doping with MnO2. Using the doping, the specific conductivity has been decreased and did not deteriorate when the samples were heated in air to elevated temperatures. Unlike in the case of N2-annealed samples, in the doped samples, saturated ferroelectric loops with negligible leakage-current contributions have been measured, revealing a coercive field of Ec~80 kV/cm and a remanent polarization of 2Pr~100 µC/cm2. Keywords: ferroelectric; BiFeO3-SrTiO3; electrical conductivity Kontrola električne prevodnosti v feroelektrični keramiki 0.7BiFeO3-0.3SrTiO3 z žganjem v dušikovi atmosferi in dopiranjem z Mn Izvleček: V tem prispevku smo z mehanokemijsko aktivacijo pripravili trdno raztopino med polarnim BiFeO3 (BFO) in nepolarnim SrTiO3 (ST) s sestavo 0.7BFO-0.3ST, pri čemer je bil poseben poudarek na karakterizaciji ter kontroli električne prevodnosti pripravljene keramike. Glede na rezultate rentgenske praškovne difrakcije in vrstične elektronske mikroskopije, vgradnja ST v BFO zmanjša koncentracijo sekundarnih faz v keramiki, ki sicer tipično nastajajo pri sintezi nemodificiranega BFO. Tako pripravljena BFO-ST keramika je pokazala visoko električno prevodnost, ki pa je bila uspešno zmanjšana z naknadnim žganjem pri 750 °C v atmosferi N2. Kljub temu je ta pristop pokazal dve pomanjkljivosti: i) ponovno oksidacijo ter posledično povečanje električne prevodnosti vzorcev, po tem ko so bili na zraku izpostavljeni povišanim temperaturam (~ 450 °C) ter ii) povečanje prevodnosti pri visokem električnem polju, kar se odraža v velikem prispevku enosmernega električnega toka k izmerjeni histerezni zanki polarizacije v odvisnosti od električnega polja. Zaradi navedenih razlogov, smo se za zmanjševanje prevodnosti odločili za alternativni pristop, tj. dopiranje z MnO2. Z uporabo tega pristopa je bila specifična prevodnost zmanjšana in se ni spremenila po segrevanju vzorcev na zraku pri povišanih temperaturah. Za razliko od vzorcev žganih v atmosferi N2, je bil v zankah dopiranih vzorcev razviden zanemarljiv prispevek prevodnosti materiala; vrednost koercitivnega polja je bila Ec ~80 kV/cm, remanentne polarizacije pa 2Pr ~100 µC/cm2. Ključne besede: feroelektrik; BiFeO3-SrTiO3; električna prevodnost * Corresponding Author’s e-mail: maja.makarovic@ijs.si M. Makarovič et al; Informacije Midem, Vol. 46, No. 3(2016), 154 – 159 1 Introduction Bismuth ferrite, BiFeO3 (BFO), is of interest due to its ferroelectric and antiferromagnetic properties at room temperature and due to its high Curie temperature (~820 °C), which make BFO a prime candidate for high-temperature piezoelectric applications [1]. The practi­cal use of BFO has, however, been limited by several fac­tors, including: i) the difficulty to prepare single-phase BFO in bulk form without the presence of secondary phases, which is related to the thermodynamic phase instability of BFO at the temperature range of ceram­ics processing, ii) the high electrical conductivity due to the mixed valence state Fe4+/Fe3+/Fe2+ and iii) the high coercive field for ferroelectric domain switching [2]. One method to suppress the formation of secondary phases, improve the electrical insulation and maintain a relatively high Curie temperature is the incorpora­tion of other ABO3 perovskites, such as PbTiO3 (PT)[3], CaTiO3 (CT)[4], BaTiO3 (BT)[5] and SrTiO3 (ST)[6], in solid solutions with BFO. Pseudo-binary BFO-ABO3 solid so­lution systems between BFO and other perovskites is also of great interest because of the possibility of in­ducing a morphotropic phase boundary (MPB) where piezoelectric properties are enhanced [7]. It has been shown in several cases that BFO-based solid solutions still exhibit high electrical conductivity. As a result, additional methods to improve the electrical insulation are required. In BFO, which is a p-type con­ductor, one such method is the annealing of the ceram­ics in an inert atmosphere with low partial pressure of oxygen, e.g., N2 [8]. Another method is chemical dop­ing with, e.g., La, Ga, Ti and Mn [5, 9–11]. In particular, Mn doping has been shown to be highly effective in improving the electrical resistivity of BFO-BT ceramics. Among the BFO-based solid solutions, the solid solu­tion between polar BFO and non-polar ST has been less investigated with little literature data regarding its electrical and electromechanical properties [6, 12, 13]. The dielectric and magnetic properties of this system as well as the potential to exhibit an MPB between a polar and non-polar phase, for which a large piezoelec­tric response has recently been predicted [14], make this solid solution system particularly interesting. The aim of this work was to prepare a BFO-ST solid so­lution in the rhombohedral (ferroelectric) region of the phase diagram reported by Fedulov [6] with the com­position containing 70 mol% of BFO (0.7BFO-0.3ST), and to measure the dielectric and ferroelectric proper­ties. Dense ceramics with minimum amount of inho­mogeneities were prepared by an alternative process­ing method, i.e., mechanochemical activation-assisted synthesis. In particular, the effect of post-annealing in nitrogen and Mn doping on the electrical conductivity was investigated. 2 Experimental work 2.1 Synthesis of the solid solution The solid solution with the composition 0.7BiFeO3-0.3SrTiO3 (0.7BFO-0.3STO) was prepared by mecha­nochemical activation-assisted synthesis using Bi2O3 (99.999%, Alfa Aesar), Fe2O3 (99.998%, Alfa Aesar), TiO2 (99.8%, Alfa Aesar) and SrCO3 (99.994% Alfa Aesar) as starting powders. After being separately milled in an absolute ethanol, the powders were weighed accord­ing to the stoichiometric ratio in a 35 g mixture and ho­mogenized in a planetary mill (Retsch PM 400, Retsch, Haan, Germany) at 200 min-1 for 4 h in absolute ethanol. The mixture was then dried and high-energy milled in a Fritch Pulverisette 7 Vario-Mill (Fritsch GmbH, Idar-Oberstein, Germany) for 40 h in an 80-ml tungsten car­bide vial, filled with 14 tungsten carbide milling balls with diameters of 10 mm. The disk rotational frequency and the vial-to-disk rotational frequency were set to, re­spectively, 300 min-1 and -2 (the negative sign denotes the opposite directions of the rotations of the vial and the disk). After the high-energy milling, the mixture was re-milled in a planetary mill in an absolute etha­nol at 200 min-1 for 4 h. Finally, the powder was dried, pressed into pellets with 150 MPa of uniaxial pressure and sintered in air at 1025 °C for 2 h using a heating/cooling rate of 5 °C/min. 2.2 Post-annealing in N2 In order to reduce the electrical conductivity of the samples, the sintered samples were post-annealed in N2 atmosphere. The samples were annealed in a nitro­gen gas flow (N2 4.6, Messer) at 750 °C for 1 h with a heating/cooling rate of 2 °C/min. 2.3 Mn doping 0.1 wt% of pre-milled MnO2 (99.9%, Alfa Aesar) was added to the activated (high-energy milled) powder. The mixture was homogenized in a planetary mill with absolute ethanol at 200 min-1 for 4 h. The powders were then pressed into pellets and sintered in air at 1025 °C for 2 h using a heating/cooling rate of 5 °C/min. 2.4 Characterization of the sintered ceramics The relative geometric density of sintered ceramics was determined using the theoretical density of 0.7BFO-0.3ST solid solution (7.40 g/cm). This theoretical den­sity was evaluated based on the theoretical density of individual perovskites, i.e., BFO (8.34 g/cm) and ST (5.12 g/cm), and their volume proportion in the solid solution. The phase composition of sintered ceramics was de­termined using X-ray powder diffraction (XRD) analysis (PANalytical X’Pert PRO diffractometer with CuK.1 ra­diation). The XRD patterns were recorded in a 2.-range from 20° to 78° with a step of 0.016° and an acquisition time of 100 s. The microstructure of sintered ceram­ics was examined by field emission scanning electron microscope (FE-SEM, JSM-7600F, Jeol) equipped with a LINK ISIS 300 (Oxford Instruments, Abingdon, U.K.) en­ergy dispersive X-ray spectrometer (EDXS). For characterization of electrical properties, the sin­tered pellets were thinned to ~0.2 mm, polished and electroded with Au by sputtering. The dielectric per­mittivity and loss tangent as a function frequency were analyzed in the range 106-10-2 Hz using a HP4284A impedance analyzer (frequency range 106-102 Hz) and Kistler charge amplifier (frequency range 103-10-2 Hz). High-electric-field polarization and strain hysteresis loops were measured by applying to the samples single sinusoidal electric-field waveforms using an aixACCT TF 2000 analyzer equipped with a laser interferometer. 3 Results and discussion 3.1 Phase composition Figure 1 shows the XRD patterns of 0.7BFO-0.3ST solid solution along with the end members of the solid so­lution (BFO and ST) for reference. The XRD pattern of 0.7BFO-0.3ST confirms the presence of the perovskite phase with no detectable secondary phases. By close inspection of the peaks of BFO and ST belonging to the {111}pc (pc denotes pseudo cubic notation) family of crystallographic planes (Figure 1 inset), it can be ob­served that BFO exhibits a pronounced {111}pc splitting to (111)pc and ( )pc peaks due to the rhombohedral unit-cell distortion, while ST exhibits a single (111) peak because of its cubic structure. Relative to BFO, BFO-ST exhibit a much weaker {111}pc peak splitting, indicat­ing that the structure is probably still rhombohedral but with a decreased lattice distortion (rhombohedral angle closer to 90°, where 90° corresponds to cubic lat­tice) With respect to BFO, all perovskite peaks of BFO-ST are shifted towards higher 2. angles, indicating a contrac­tion in unit cell volume by the addition of ST to BFO. Figure 1: XRD patterns of BFO, ST and 0.7BFO-0.3ST solid solution. The inset on the right side of the figure shows an enlarged view of the peaks belonging to the {111}pc family of crystallographic planes. 3.2 Microstructure The microstructure and phase composition of the sin­tered ceramics were investigated by FE-SEM (Figure 2). The ceramics exhibit a dense microstructure, consis­tent with the measured relative density (.rel) of ~97%. The chemical homogeneity of the rhombohedral perovskite matrix phase was investigated using EDXS analysis performed on 14 randomly selected points. The standard deviations calculated over all measure­ments for Bi, Fe, Sr and Ti were 0.7, 1.2, 1.9 and 2.1 %, re­spectively. The uncertainty of a standard-less EDS anal­ysis was reported to be ±5% relative [15]. These results show that the deviations within the analyzed sample are smaller than the uncertainty of the method, indi­cating the homogeneous distribution of the elements. However, a closer inspection of the microstructure (see the inset in Figure 2) revealed small, submicron sized chemically inhomogeneous inclusions, the concentra­tion of which is evidently insufficient to be detected by XRD, since the ceramics was identified as single phase using XRD analysis (Figure 1). 3.3 Dielectric properties The real part of the complex dielectric permittivity (.'), the dielectric loss tangent (tan.) and the real part of the complex electrical conductivity (.') were studied as a function of frequency for the 0.7BFO-0.3ST ceramics. The real part of the electrical conductivity .' for each frequency data point was calculated from the mea­sured imaginary dielectric permittivity (.d’’) and fre­quency w using equation (1): (1) where .0 is the permittivity of vacuum. Four different samples have been analyzed: i) as-sin­tered (undoped) ceramics (.), which was ii) post-an­nealed in N2 at 750 °C (0), and iii) additionally annealed in air at ~450 °C after the N2 annealing (.) and iv) Mn-doped ceramics (.). The as-sintered ceramics (Fig. 3, .) exhibit two relax­ations in the measured frequency range, observed as two step-like features in .' within the frequency range 101-105 Hz, which are accompanied by two peaks in tan.. These relaxations result in high apparent .' values at the low frequency limit (.'>105 below 10-2 Hz). Such relaxation behavior in BFO has been earlier attributed to either Maxwell Wagner-like or polaron hopping con­duction mechanisms [2], both in principle related to the elevated electrical conductivity of the ceramics. After the sintered 0.7BFO-0.3STceramics were annealed in N2, the dielectric relaxations were largely suppressed (Fig. 3, 0). From the frequency dependent .', it is pos­sible to estimate the specific electrical conductivity (.0) using further expansion of equation 1 into equation 2: (2) where .d’’ is the imaginary part of the dielectric per­mittivity related to dielectric (or polarization) losses. At low frequencies, if the term ..0.d’’ is sufficiently small, .' in principle reaches a plateau and .0 can be estimated as .' . .0. Even though this plateau in .' has not been reached in the as-sintered (Fig. 3c, .) and N2-annealed (Fig. 3c, 0) samples, the lower values of .' of the last-mentioned sample in the low-frequency range indicate a reduction of the electrical conductivity, pos­sibly related to the reduction of Fe4+ to Fe3+ ions during N2-annealing as earlier reported for BFO. [8] However, after the same sample was annealed at ~450°C in air, the dielectric relaxation was again observed. Therefore, the results suggest that the sample re-oxidizes during annealing in air, meaning that the annealing in N2 has limited practical implications in reducing efficiently the electrical conductivity of BFO-ST. Similar as in the case of N2-annealing, Mn doping sup­pressed the dielectric relaxation originally observed in the as-sintered and undoped sample, resulting in a similar .', tan. and .'. In a strong contrast, however, the electrical conductivity and the associated relaxation of this doped composition did not deteriorate after the sample was heated in air at ~450°C (not shown). Figure 3: a) Room-temperature real part of complex di­electric permittivity (.'), b) dielectric loss tangent (tan.) and c) real part of complex electrical conductivity (.') as a function of frequency for 0.7BFO-0.3ST as-sintered ceramics (.), annealed in N2 at 750 °C (0) and addition­ally annealed in air at 450°C (.), and doped with MnO2 (.). 3.4 Ferroelectric properties The polarization-electric-field (P-E) hysteresis loops of 0.7BFO-0.3ST ceramics annealed in N2 and doped with Mn are shown in Figure 4. The loops were measured at room temperature at 120 kV/cm and frequency of 100 Hz and 10 Hz for the annealed and the doped sample, respectively. The test frequencies were different be­cause the N2-annealed ceramics experienced high leakage currents and dielectric breakdown at test fre­quencies below 100 Hz. Despite the difference in the test frequency, the comparison of the loops can still be used to achieve a qualitative assessment of the differ­ent high electric-field behavior of each ceramic. The P-E loop of the N2-annealed sample is not saturated and shows high leakage-current contribution, evidenced by the rounded shape of the loop. On the contrary, the P-E loop of Mn-doped sample shows approximately saturated loops, signified by the sharp corners at the maximum and minimum electric fields, with the coer­cive field Ec ~80 kV/cm and remanent polarization 2Pr ~100 µC/cm2. The inset on the right side of the figure shows the loops for the corresponding samples at 10 kV/cm. By comparing the loops of the N2 annealed sample at 10 kV/cm and 120 kV/cm it can be observed, that there is an abrupt increase in conductivity at high electrical fields, which results in strong leakage-current contribution to the measured polarization-electric-field loops (rounded loop in Fig. 4). Figure 4: Room temperature P-E loops of 0.7BFO-0.3ST ceramics annealed in N2 and Mn doped measured at 120 kV/cm. The inset on the right side shows P-E loops of the same samples measured at 10 kV/cm. 4 Conclusions In the present report, ceramics with composition 0.7BFO-0.3ST were prepared by mechanochemical activation-assisted synthesis, with high chemical ho­mogeneity and high bulk density. The as-sintered samples exhibited high electrical conductivity, which was successfully suppressed by i) post-annealing of the sintered samples in a nitrogen flow or ii) doping with Mn. However, the exposure of the annealed samples to elevated temperature in air resulted in re-oxidation of the samples and consequently re-gain of the high electrical conductivity. In addition, the P-E loops of post-annealed samples showed increased conductiv­ity at high electrical fields, resulting in leakage-current dominated P-E loops. In contrast to the post-annealing in N2 atmosphere, the doping with Mn has been proven as an efficient method to reduce the electrical conduc­tivity in this system. As expected, the conductivity of Mn-doped samples did not deteriorate when heated in air to elevated temperatures. The P-E loops were satu­rated with Ec ~80 kV/cm and 2Pr ~100µC/cm2. 5 Acknowledgments This work was supported by the Slovenian Research Agency (program P2-0105 and project J2-5483). 6 References 1. G. Catalan and J. F. Scott, “Physics and Applica­tions of Bismuth Ferrite,” Adv. Mater., vol. 21, no. 24, pp. 2463–2485, Jun. 2009. 2. T. Rojac, A. Bencan, B. Malic, G. Tutuncu, J. L. Jones, J. E. Daniels, and D. Damjanovic, “BiFeO3 Ceram­ics: Processing, Electrical, and Electromechanical Properties,” J. Am. Ceram. Soc., vol. 97, no. 7, pp. 1993–2011, Jul. 2014. 3. T. P. Comyn, S. P. McBride, and a. J. Bell, “Processing and electrical properties of BiFeO3–PbTiO3 ceram­ics,” Mater. 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