ISSN 0352-9045 Journal of Microelectronics, Electronic Components and Materials Vol. 52, No. 4(2022), December 2022 Revija za mikroelektroniko, elektronske sestavne dele in materiale letnik 52, številka 4(2022), December 2022 UDK 621.3:(53+54+621+66)(05)(497.1)=00 ISSN 0352-9045 Informacije MIDEM 4-2022 Journal of Microelectronics, Electronic Components and Materials VOLUME 52, NO. 4(184), LJUBLJANA, DECEMBER 2022 | LETNIK 52, NO. 4(184), LJUBLJANA, DECEMBER 2022 Published quarterly (March, June, September, December) by Society for Microelectronics, Electronic Components and Materials - MIDEM. Copyright © 2022. All rights reserved. | Revija izhaja trimesecno (marec, junij, september, december). Izdaja Strokovno društvo za mikroelektroniko, elektronske sestavne dele in materiale – Društvo MIDEM. Copyright © 2022. Vse pravice pridržane. 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Prispevke iz revije zajema ISI® v naslednje svoje produkte: Sci Search®, Research Alert® in Materials Science Citation Index™. Design | Oblikovanje: Snežana Madic Lešnik; Printed by | tisk: Biro M, Ljubljana; Circulation | Naklada: 1000 issues | izvodov; Slovenia Taxe Percue | Poštnina placana pri pošti 1102 Ljubljana Electronic Components and Materials vol. 52, No. 4(2022) Content | Vsebina Original scientific papers Izvirni znanstveni clanki R. Fathima, S. Perumal, V. Muniyappan, M. Faseehuddin, W. Tangsrirat: Electronically Tunable Multifunction Current Mode Filter Employing Grounded Capacitors 205 R. Fathima, S. Perumal, V. Muniyappan, M. Faseehuddin, W. Tangsrirat: Elektronsko nastavljiv vecfunkcijski fi lter v tokovnem nacinu z ozemljenimi kondenzatorji S. Bernik, N. Brguljan, M. Ercegovac, Z. Samardžija: Influence of Trace Eelements on the Electrical Properties of ZnO-based Multilayer Varistors 215 S. Bernik, N. Brguljan, M. Ercegovac, Z. Samardžija: Vpliv elementov v sledovih na elektricne lastnosti vecplastnih varistorjev na osnovi ZnO R. Mishra, G. R. Mishra, M. Faseehuddin, J. Sampe: VD-EXCCII Based Mixed Mode Biquadratic Universal Filter EmployingGrounded Capacitors 227 R. Mishra, G. R. Mishra, M. Faseehuddin, J. Sampe: VD-EXCCII na osnovi mešanega nacina bikvadraticnega univerzalnega filtra, ki uporablja ozemljene kondenzatorje D. Singh, S. K. Paul: Mixed-mode Universal Filter Using FD-CCCTA and its Extension as Shadow Filter 239 D. Singh, S. K. Paul: Univerzalni filter z mešanim nacinom uporabe FD-CCCTA in njegova razširitev kot filter v senci D. Palani, M. Arulraj:: User Offloading using Hybrid NOMA in Next-generation Heterogeneous Network 263 D. Palani, M. Arulraj: Uporabniška razbremenitev z uporabo hibridnega NOMA v heterogenem omrežju naslednje generacije Winners of Prestigious International Awards 271 Dobitnika uglednih mednarodnih nagrad Announcement and Call for Papers: 58th International Conference on Microelectronics, Devices and Materials With the Workshop on Chemical sensors: materials and applications 273 Napoved in vabilo k udeležbi: 58. mednarodna konferenca o mikroelektroniki, napravah in materialih z delavnico o kemicnih senzorjih: materialih in aplikacijah Front page: SEM image showing typical morphology of the CrO powders (S. Bernik et al.) 23 Naslovnica: SEM slika tipicne morfologije praška CrO23 (S. Bernik et al.) Original scientific paper https://doi.org/10.33180/InfMIDEM2022.401 Electronic Components and Materials Vol. 52, No. 4(2022), 205 – 214 Electronically Tunable Multifunction Current Mode Filter Employing Grounded Capacitors Rani Fathima1, Srideviponmalar Perumal2, Vadivel Muniyappan3, Mohammad Faseehuddin4, Worapong Tangsrirat5 1Electrical Engineering Program, EDICT Department, Bahrain Polytechnic, Bahrain. 2Department of Computational Intelligence, School of computing, SRM Institute of Science and Technology, India 3Department of Electronics and Communication, Vidya Jyoti Institute of Technology, India 4Department of Electronics and Telcommunication, Symbiosis Institute of Technology (SIT), Symbiosis International (Deemed) University (SIU), India 5Department of Instrumentation and Control Engineering, School of Engineering, King Mongkut’s Institute of Technology Ladkrabang (KMITL), Bangkok, Thailand Abstract: In this paper, a new single input multi output (SIMO) filter is presented that works in current mode (CM). The universal filter is designed using a recently proposed highly versatile active building block the extra X current conveyor transconductance amplifier (EXCCTA). The design employs two EXCCTAs, two capacitors and one resistor. The designed filter uses grounded passive elements which is advantageous for fabrication. The design can provide all the five responses i.e., high-pass (HP), band-pass (BP), low-pass (LP), all-pass (AP), and band-stop (BS) simultaneously. In addition, it provides an independent electronic tunability of angular frequency (.) and quality factor (Q). Moreover, there is no requirement of passive component matching. The non-ideal and sensitivity analysis of the filter is done to get a measure of the effect of the process and components variation on the functioning of the filter. The simulation results are obtained using Cadence software employing 0.18 µm CMOS technology parameters from Silterra Malaysia at a supply voltage of ±1.25 V also the layout of the EXCCTA is designed. The proposed filter is validated by designing it for a frequency of 16. 07MHz. Additionally, the Spice macro model of the commercially available integrated circuits (ICs) AD844 and LM13700 are used to further test the feasibility of the proposed filter. Keywords: current mode; filter; current conveyor; universal filter; analog Elektronsko nastavljiv vecfunkcijski filter v tokovnem nacinu z ozemljenimi kondenzatorji Izvlecek: V clanku je predstavljen nov enovhodni vecizhodni filter (SIMO), ki deluje v tokovnem nacinu (CM). Univerzalni filter je zasnovan z uporabo nedavno predlaganega zelo vsestranskega aktivnega gradnika - transkondukcijskega ojacevalnika z dodatnim tokom X (EXCCTA). Zasnova uporablja dva EXCCTA, dva kondenzatorja in en upor. Zasnovani filter uporablja ozemljene pasivne elemente, kar je ugodno za izdelavo. Zasnova lahko hkrati zagotavlja vseh pet odzivov, tj. visokoprepustni (HP), pasovni (BP), nizkoprepustni (LP), vseprepustni (AP) in pasovno zaporo (BS). Poleg tega omogoca neodvisno elektronsko nastavitev kotne frekvence (.) in faktorja kakovosti (Q). Poleg tega ni potrebe po usklajevanju pasivnih komponent. Vplivi sprememb procesa in komponent na delovanje filtra so bili izmerjeni s pomocjo neidealne analize in analize obcutljivosti filtra. Rezultati simulacije so pridobljeni s programsko opremo Cadence, pri cemer so uporabljeni parametri tehnologije CMOS 0,18 µm podjetja Silterra Malaysia pri napajalni napetosti ±1,25 V. Zasnovana je tudi postavitev EXCCTA. Predlagani filter je potrjen za frekvenco 16,07 MHz. Poleg tega sta za nadaljnje testiranje izvedljivosti predlaganega filtra uporabjena makro modela Spice komercialno dostopnih integriranih vezij (IC) AD844 in LM13700. Kljucne besede: tokovni nacin; filter; tokovni transporter; univerzalni filter; analogni * Corresponding Author’s e-mail: sridevip@srmist.edu.in How to cite: R. Fathima et al., “Electronically Tunable Multifunction Current Mode Filter Employing Grounded Capacitors", Inf. Midem-J. Microelectron. Electron. Compon. Mater., Vol. 52, No. 4(2022), pp. 205–214 R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 1 Introduction From the last few decades, the designing of current mode (CM) analog filters has gained popularity among researchers due to their versatility and wide applicabil­ity. Their applications can be easily found in high-speed communication, instrumentation, sound system, con­trol engineering, and electroacoustic etc.[1-4]. Presently universal filters designed using low voltage low power (LVLP) techniques are in demand because of the emer­gence of portable battery-operated devices. A univer­sal filter circuit provides all the five filter responses, i.e. high-pass (HP), low-pass (LP), band-pass (BP), band-stop (BS), and all-pass (AP), from the same topology[3]. Fur­thermore, universal filters can be categorized as single input multi output (SIMO)[1, 3], multi-input multi out­put (MIMO)[1, 5] and multi input single output (MISO) [6, 7] filters. Second order filters have wider range of applications, so their design is an important area of re­search. Considering the benefits current mode (CM) cir­cuits have in terms of higher bandwidth, good dynamic range and low power dissipation, the proposed univer­sal filter is designed using the CM active block. Several SIMO universal filters were designed employing differ­ent CM active blocks by researchers in the literature[2, 4, 5, 8-30]. Some of these active blocks are differential voltage current conveyor (DVCC) [2, 8], current conveyor transconductance amplifier (CCTA) [9], current follower transconductance amplifier (CFTA) [11], operational floating current conveyor (OFCC) [24], third generation Table 1: Comparative studey of the CM SIMO universal filters current conveyor (CCIII)[10], second generation current conveyor (DOCCII) [8, 9, 13], four terminal floating nul­lor transconductance amplifier (FTFNTA) [21], and extra x current conveyor transconductance amplifier[27], volt­age differencing current conveyor. A comparative study of some exemplary designs of CM SIMO filters is done based on the following parameters (i) Number of analog building blocks required (ABBs) (ii) Number of Passive Components employed (iii) Grounded passive compo­nents used in the design (iv) the filter has low input im­pedance (v) all responses are available through explicit high impedance terminals (vi) responses available (vii) electronic tunability feature present (viii) independent control of quality factor and pole frequency (ix) design frequency. The Table 1 presents the comparative analy­sis. The available designs have some limitations in terms of cascading feature, number of passive elements, num­ber of floating passive components, independent tun-ability of frequency and quality factor and simultaneous availability of all five filter responses as mentioned below. - Low output impedance due to which cascading is not possible [9, 10, 26, 27]. - High input impedance which is undesirable for cascading [8-10, 12, 18, 14-26, 30]. - More than two active elements are employed for the design [8, 11, 12, 13, 15, 17, 19, 28, 30]. -Angular frequency and quality factor are not inde­pendently tunable [8-10, 12, 16, 17, 19, 24, 26-28]. - Fabrication is difficult due to the availability of floating passive elements [10, 12]. References (i) (ii) (iii) (iv) (v) (vi) (vii) (viii) (ix) [8] DVCC (3) 4R+2C Yes No Yes All fi ve No No 22.5MHz [9] CCTA (1) 2R+2C Yes No No All fi ve Yes No 1MHz [10] CCIII (1) 2R+2C No No No LP, BP No No 562.7kHz [11] CFTA (4) 2C Yes Yes Yes All fi ve Yes Yes 153kHz [12] MOCCII (3) 5R+2C No No Yes All fi ve No No 281.35kHz [13] CCII (3) 3R+2C Yes Yes Yes All fi ve No Yes 1MHz [15] ZC-CFTA (4) 2C Yes Yes Yes All fi ve Yes Yes 159kHz [16] ZC-CITA (2) 2C Yes Yes Yes All fi ve Yes No 1.026MHz [17] MOCCII (3) 2R+2C Yes Yes Yes All fi ve No No 1kHz [18] VDCC (2) 2R+2C Yes No Yes All fi ve Yes Yes 1.06MHz [19] MOCCII (3) 2R+2C Yes Yes Yes All fi ve No No 436.2kHz [24] MO-OFC (2) 2R+2C Yes No Yes All fi ve No No 1.5MHz [25] DXMOCCII (2) 2R+2C Yes No Yes All fi ve Yes Yes 2.65MHz [25] DXMOCCII (2) 1R+2C Yes No Yes All fi ve Yes Yes 2.65MHz [26] VDCC (1) 2R+2C Yes No No All fi ve Yes No 8.91MHz [27] EXCCTA (1) 1R+2C Yes Yes No All fi ve Yes No 2.054MHz [28] CFTA (3) 2C Yes Yes Yes All fi ve Yes No 6.4MHz [29] DXMOCCII (2) 3R+2C Yes Yes Yes All fi ve No Yes 1.203MHz [30] MOCCII (3) 3R+2C Yes No Yes All fi ve No Yes 100MHz Proposed EXCCTA (2) 1R+2C Yes Yes Yes All fi ve Yes Yes 16.07MHz R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 - All five responses of filters are missing [10]. - Capacitor is connected to low impedance node which will degrade high frequency performance [25]. This paper describes the design of a SIMO filter by mak­ing use of two EXCCTA, two grounded capacitors and one resistor. It provides all five responses concurrently and it features independent control of angular frequen­cy (.) and quality factor (Q) via transconductance of the EXCCTA. Another important advantage of the filter is that the outputs are available explicitly from high impedance terminals which are essential for cascading point of view. The design is validated using Cadence design software and the simulations results are found to be in closely fol­lowing the expected theoretical results. 2 Extra X Current Conveyor Transconductance Amplifier (EXCCTA) The Extra X current conveyor transconductance am­plifier (EXCCTA) is functionally an improved and more versatile version of extra x current conveyor (EXCCII) [31]. The EXCCTA[27] includes features of current and voltage followers and operational transconductance amplifier (OTA) making it more versatile. The voltage current (V-I) characteristics of the developed EXCCTA are given in Equations (1-5) and the block diagram is presented in Figure 1. Figure 1: Block Diagram of EXCCTA V  V V (1) XP XN Y I  I I (2) XP ZP  ZP  I  I I (3) XN ZN  ZN  I g  V  (4) O  m ZP  The expression for transconductance (gm) is given in Equation 5. W g  µ CI (5) m nOX B L Where CQX is the gate oxide capacitance, µn is the mobil­ity of electrons in NMOS, gm denotes the transconduct­ance of OTA set via bias current IB and W/L is the aspect ratio of the transistors. The CMOS implementation of the EXCCTA as proposed in[27] is presented in Figure 2. The Y terminal is high impedance voltage input node and XP & XN low imped­ance voltage output/current input nodes. The O+, ZP+ & ZN+ terminals are high impedance current output nodes. The number of current output terminals (I, I ZP+ZP-, I, I, O, O) can be increased by simply adding two ZN+ZN-+­ MOS transistors. 3 Proposed EXCCTA based CM SIMO filter The proposed current mode SIMO filter is shown in Fig­ure 2. It employs two EXCCTA, one grounded resistor and two grounded capacitors which is advantageous for fabrication point of view. The filter is fully cascad-able having low input impedance and high output im­pedance. Additionally, the pole frequency and quality factor of the filter can be independently tuned via bias current of the OTA. Another important design feature Figure 2: CMOS implementation of EXCCTA R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 is the use of only positive current output terminals as it avoids the use of additional MOS transistors for current reversal and improves accuracy. Figure 3: Proposed SIMO universal filter The analysis of the filter circuit yields the transfer func­tions of all the five filter responses as given in Equations (6-10). The expression of quality factor and pole fre­quency of the filter are presented in Equations (11-13). I HP I IN  2 S C C R 121 2 S C C R  SC R g 121 11m2  g m1 (6) I LP I IN  2 S C C R 121 g m1  SC R g 11m2  g m1 (7) I BP I IN  SC R g 11m2 2 S C C R  SC R g 121 11m2  g m1 (8) I NP I IN  2  SC C R  g 121 m1 2 S C C R  SC R g 121 11m2 g m1 (9) I AP I IN  2  S C C R  g  SC R gm 121 m1 11 2 2 S C C R  SC R gm g 121 11 2 m1 (10) f o  1 g m1 2 CC R 121 (11) Q  1 gC m12 g C R m2 11 (12) BW  1 g m 2 2 C 2 (13) 2 11 PP 1 N From equation (11) to (12), it is very clear that we can independently tune the quality factor of the filter with­out affecting the frequency (f) which means that f and Q are orthogonally tunable. 4 Non - ideal and sensitivity analysis The imperfections present in the MOS transistors leads to improper transfer of current and voltage signals which leads to a shift in the V-I transfer characteristics of the EXCCTA from the ideal one. This results in the shift in the frequency and quality factor of the designed filter. The frequency dependent current, voltage and transconductance transfer gains are considered for the analysis as they are the major contributor. Considering the non-ideal gains the V-I relations of the EXCCTA will be modified to IY=0, V = ßp(s)VY, V = ßN(s)VY, IZP+ = a (s) XPXNp I = a(s)I = .gV, where ß is non-ideal volt- XP, IZN+NXN, I0+m ZP+P/N age transfer gain, aP/N is non-ideal current transfer gain and . is non-ideal transconductance transfer gain. Ide­ally ß = a = . = 1. P/NP/N By considering the effect of EXCCTA non-idealities on the designed filter the expression of quality factor and angu­lar frequency are modified as given in Equations 14-15.  g 1 NPP m1 (14) fo 2 CC R 121 1  gC NP m12 (15) Q g  CR m2 P 11 The active and passive sensitivities of the proposed are evaluated and presented below.   R  1 SC SC SSS  S  S  Sg , 121 PNP 1 2 1 Q QQQQ QQQQ S  S  S  S  S  S  SS  S  CCR  g  2 Q Sg  1 2 It is clear from analysis that all the sensitivities are unity or below which is the required condition. Hence the proposed filter has good performance in terms of sen­sitivity. 5 Parasitic study The effect of EXCCTA parasitic elements on the per­formance of the filter is carried out in this section. The EXCCTA parasitic elements are shown in Figure 4. At Y terminal the parasitic resistance and capacitance ap­pear in parallel (RY\\CY), same is the case with parasitic elements at ZP, ZN and O+ terminals where the parasitic elements appear in parallel as follows: (R\\C), (R\\ ZPZPZN R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 C), and (R\\C). The parasitic associated with low ZNO+O+ impedance XP and XN terminals appear as a resistors RXP and RXN in series with an inductor. The inductive effect is dominant at a very high frequency so it is ignored in this analysis. Adding the EXCCTA parasitic elements in the proposed filter the node capacitance and resistance will me mod­ified as C’ = (C\\C\\C\\C), C’ = (C\\C), R’ = (R\\ 22YZP1+O+11ZP1+11 RXN). The modified tranfer functions of the filter includ­ing the parasitics are presented in Equations 16-22. The change in the total node capacitance and resistance majorly result in the deviation.One advantage of this topology is that the capacitors are connected with the high impedance nodes and the resistor is connected with the low impedance node. The capacitors will ab­sorb the parasitic capacitance and the connected re­sistance will absorb the parasitic series resistance they by reducing the effect on the performance. 2''' I S RCC HP 11 2  (16) 2''' '' I S RCC  SRC g g IN 112 11m2 m1 Ig LP m1  (17) 2''' '' I S RCC  SRC g g IN 112 11m2 m1 '' I SRC g BP 11m2  (18) 2''' '' I S RCC  SRC g g IN 112 11m2 m1 2'' ' I  SRCC  g NP 121 m1  (19) 2''' '' I S RCC  SRC g g IN 112 11m2 m1 2'' ' '' I  S RCC  g  SRC gm AP 121 m1 11 2  (20) 2''' '' I S RCC  SRC g g IN 112 11m2 m1 1 g m1 f  (21) o ''' 2 RCC 112 ' 1 gC m1 2 Q  (22) '' g RC m2 11 6 Simulation results To validate the proposed resistor less CM SIMO filter it is designed and simulated in Cadence virtuoso design software. The EXCCTA is designed in 0.18 µm Silterra Malaysia technology at a supply voltage of ±1.25V. The width and length of the transistors used are given in Figure 4: Non-ideal equivalent circuit model of the EX­CCTA Table 2. The transconductance of the OTA was fixed 1.02 mS by selection the bias current Ibias = 120 µA. The layout of the EXCCTA is presented in Fig. 5. It is drawn using the nhp and php high performance MOS transis­tors from the Silterra library and occupies chip area of 65*26µm2. Table 2: Width and Length of the MOS transistors Transistors Width (µm) Length (µm) M1-M2, M5-M6 1.8 0.36 M3, M4, M7, M8, M9 5.4 0.36 M10-M14 1.8 0.72 M15-M18 3.06 0.36 M19-M22 10 0.36 M23, M25, M27, M29, M31, M35, M37, M33, M42, M44 2.16 0.36 M24, M25, M28, M41, M43, M34 0.72 0.72 M30, M32, M36, M38 1.08 0.72 The pole frequency of the filter is fixed at 16.07 MHz and quality factor to 1.2 by setting passive component values as R1 = 1kO, C1 = C2 = 10 pF and gm = 1.02 µS. The LP, HP, BP and NP responses of the CM SIMO filter are presented in Fig. 6. The AP gain and phase response is given in Fig. 7. The simulated frequency for CM-AP is found to be 15.75 MHz leading to 2% error. The power dissipation of the filter is found to be 3.29mW. R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 Figure 5: Layout of the EXCCTA be seen the filter output is accurate in terms of phase and magnitude. The Monte Carlo analysis is performed to measure the effect of device parameter variations and process spread on the performance of the filter. The Monte Carlo analysis is done for 200 runs using the mismatch models provided in the PDK for MOSFETs. As can be seen from Figure 8 the mean frequency is found to be 15.79 MHz which is close to designed frequency. Also, the frequency spread is small for majority of the samples so it can be concluded that the filter exhibits acceptable performance with minimum deviation un­der process variations. It can also be seen from the plot in Fig. 9 that minimum and the maximum frequency are 14.9MHz and 16.81MHz respectively. The phase plot of the all-pass response for the Monte Carlo analy­sis is presented in Fig. 10. R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 ranging from I =20µA(g = 463µS), I =50µA(g = Bias2m2Bias2m2 681µS), I =80µA(g = 910µS) and I =120µA(g = Bias2m2Bias2m2 1.02µS) while keeping I constant as presented in Fig. Bias1 11. The quality factor of the filter chages as per Equa­tion 12 due to the change in the transcoducatnnce (gm2). The effect of temperature variation on the func­tioning of the filter is examined by analysing the BP response under different temperature values ranging from 0o to 1000C. It can be inferred for the graph in Fig. 12(a-b) that although the filter frequency decreases with increase in temperature it is close to theoretical value within 20o to 600C. Theoretically, the frequency of the filter should not change with temperature, but the frequency of the filter decreases due to rise in tempera­ture because of the decrease in OTA transconductance (gm). Two main contributing factors that influence the transconductance are the threshold voltage (Vt) and carrier mobility.The total harmonic distortion (THD) of the filter is measured for different input current ampli­tudes as presented in Fig. 13. The THD remains within acceptable limit of 5% for significant input current range. Figure 11: BP quality factor tuning (a) (b) R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 The frequency tunability of the filter is shown by vary­ing the bias currents (Iand I) of the OTAs simulta­ Bias1 Bias2 neously. It can be seen from Figure 14 that the filter fre­quency changes with change in the bias currentss also it is observed that there is slight change in the quality factor of filter due to change in the transconductnace (gm2). The change in the quality factor can be nullified by according­ly setting the value of R or setting the value of g 1m2. 7 Implementation using commercially ICs To further investigate the workability of the filter it is designed using commercially available ICs. The be- Figure 14: Variation of filter frequency with bias cur­rents of the OTA havioural models of the current feedback operational amplifier (AD844) and OTA (LM13700) provided by the manufacturer are used in the study. The PSPICE soft­ware is used for the analysis. The authors want to men­tion that the IC based implementation is done for the proof of concept as the fabricated chip is not available for experimental validation. The Figure 15 presents the implementation of the filter. The filter frequency is set at 225kHz by setting OTA transconductance at 2mS and passive component val­ues at R1 = 1 kO, C1 = C2 = 1nF. The frequency domain analysis of the filter is done as presented in Figure 16. The close relationship between the ideal and simulated results verify the feasibility of the proposed design. R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 8 Conclusion This paper presents a new EXCCTA based electronically tunable SIMO filter. The filter employs two EXCCTA, one resistor and two grounded capacitors. Presented SIMO filter has inbuilt tunability and can realize all five filter responses simultaneously. The EXCCTA is designed in Cadence Virtuoso software and extensive simulations are carried out to examine and validate the proposed filter. The proposed filter has all the advantages men­tioned in Table 1.The filter is designed for a frequency of 16.07 MHz at ±1.25 V supply. The Monte Carlo analy­sis shows that the frequency deviation is within ac­ceptable limits. Furthermore, the THD is within 5% for considerable current input signal range. The simulation results are found consistent with the theoretical predic­tions. 9 Conflicts of interest The authors declare no conflict of interest. 10 Acknowledgement This work was supported by King Mongkut’s Institute of Technology Ladkrabang (KMITL). 11 References 1. R. Senani, D. Bhaskar, and A. Singh, Current con­veyors: variants, applications and hardware imple­ mentations vol. 560: Springer, 2015. https://doi.org/10.1007/978-3-319-08684-2. 2. J.-W. Horng and Z.-Y. Jhao, “Voltage-mode univer­sal biquadratic filter using single DVCC,” Interna­tional Scholarly Research Notices, vol. 2013, 2013. https://doi.org/10.1155/2013/125746. 3. P. A. Mohan, Current-mode VLSI analog filters: de­sign and applications: Springer Science & Business Media, 2003. https://doi.org/10.1007/978-1-4612-0033-8. 4. A. M. Soliman, “Current-mode universal fi lters us­ing current conveyors: classification and review,” Circuits, Systems & Signal Processing, vol. 27, pp. 405-427, 2008. https://doi.org/10.1007/s00034-008-9034-y 5. S. V. Singh, S. Maheshwari, and D. Chauhan, “Cur­rent-processing current-controlled universal bi-quad fi lter,” Radioengineering, vol. 21, pp. 317-323, 2012. https://doi.org/10.1007/s00034-008-9034-y. 6. H.-P. Chen, “Versatile current-mode universal bi-quadratic filter using DO-CCIIs,” International jour­nal of electronics, vol. 100, pp. 1010-1031, 2013. https://doi.org/10.1080/00207217.2012.731370. 7. H. P. Chen, “Current-mode dual-output ICCII-based tunable universal biquadratic fi lter with low-input and high-output impedances,” Interna­tional Journal of Circuit Theory and Applications, vol. 42, pp. 376-393, 2014. https://doi.org/10.1002/cta.1858. 8. M. A. Ibrahim, S. Minaei, and H. Kuntman, “A 22.5 MHz current-mode KHN-biquad using diff erential voltage current conveyor and grounded passive elements,” AEU-International Journal of Electronics and Communications, vol. 59, pp. 311-318, 2005. https://doi.org/10.1016/j.aeue.2004.11.027. 9. N. Herencsar, J. Koton, and K. Vrba, “Single CCTA-Based Universal BiquadraticFilters Employing Minimum Components,” International Journal of Computer and Electrical Engineering, vol. 1, p. 307, 2009. https://doi.org/10.7763/IJCEE.2009.V1.48. 10. E. Yuce, B. Metin, and O. Cicekoglu, “Current-mode biquadratic filters using single CCIII and mini­mum number of passive elements,” Frequenz, vol. 58, pp. 225-227, 2004. https://doi.org/10.1515/FREQ.2004.58.9-10.225. 11. W. Tangsrirat, “Single-input three-output elec­tronically tunable universal current-mode fi lter using current follower transconductance ampli­fi ers,” AEU-International Journal of Electronics and Communications, vol. 65, pp. 783-787, 2011. https://doi.org/10.1016/j.aeue.2011.01.002. 12. J.-W. Horng, “Current-mode and transimpedance-mode universal biquadratic filter using multiple outputs CCIIs,” 2010. R. Fathima et al.; Informacije Midem, Vol. 52, No. 4(2022), 205 – 214 13. K. K. Abdalla, D. R. Bhaskar, and R. Senani, “Con­figuration for realising a current-mode universal filter and dual-mode quadrature single resistor controlled oscillator,” IET circuits, devices & systems, vol. 6, pp. 159-167, 2012. https://doi.org/10.1049/iet-cds.2011.0160. 14. A. Qadir and T. Altaf, “Current mode canonic OTA­C universal filter with single input and multiple outputs,” in 2010 2nd International Conference on Electronic Computer Technology, 2010, pp. 32-34. https://doi.org/10.1109/ICECTECH.2010.5479995. 15. J. Satansup and W. Tangsrirat, “Single-input fi ve-output electronically tunable current-mode bi-quad consisting of only ZC-CFTAs and grounded capacitors,”Radioengineering, vol. 20, pp. 650-655, 2011. 16. D. Biolek, V. Biolkova, Z. Kolka, and J. Bajer, “Sin-gle-input multi-output resistorless current-mode biquad,” in 2009 European Conference on Circuit Theory and Design, 2009, pp. 225-228. https://doi.org/10.1109/ECCTD.2009.5274928. 17. R. Senani and A. Singh, “A new universal current-mode biquad fi lter,” Frequenz, vol. 56, pp. 55-59, 2002. https://doi.org/10.1515/FREQ.2002.56.1 -2.55. 18. M. Gupta, P. Dogra, and T. S. Arora, “Novel current mode universal filter and dual-mode quadrature oscillator using VDCC and all grounded passive elements,” Australian Journal of Electrical and Elec­tronics Engineering, vol. 16, pp. 220-236, 2019. https://doi.org/10.1080/1448837X.2019.1648134. 19. M. Sagbas and M. Koksal, “Current-mode state-variable fi lter,” Frequenz, vol. 62, pp. 37-42, 2008. https://doi.org/10.1515/FREQ.2008.62.1 -2.37. 20. J. Jerabek, R. Sotner, and K. Vrba, “Comparison of the SITO current-mode universal fi lters using multiple-output current followers,” in 2012 35th International Conference on Telecommunications and Signal Processing (TSP), 2012, pp. 406-410. https://doi.org/10.1109/TSP.2012.6256325. 21. R. Singh and D. Prasad, “Electronically Tunable SIMO type Mixed-mode Biquadratic Filter using Single FTFNTA,” Indian Journal of Pure & Applied Physics (IJPAP), vol. 59, pp. 629-637, 2021. 22. L. Safari, G. Barile, G. Ferri, and V. Stornelli, “A new low-voltage low-power dual-mode VCII-based SIMO universal fi lter,” Electronics, vol. 8, p. 765, 2019. https://doi.org/10.3390/electronics8070765. 23. A. Abaci and E. Yuce, “A new DVCC+ based sec-ond-order current-mode universal fi lter consist­ing of only grounded capacitors,” Journal of Cir­cuits, Systems and Computers, vol. 26, p. 1750130, 2017. https://doi.org/10.1142/S0218126617501304. 24. T. Parveen, “OFC based high output impedance current mode simo universal biquadratic fi lter,” in 2011 International Conference on Multimedia, Sig­nal Processing and Communication Technologies, 2011, pp. 134-136. https://doi.org/10.1109/MSPCT.2011.6150456. 25. A. Kumar, A. K. Kushwaha, and N. Chander, “Cur-rent-mode SIMO universal filter realization using dual-X multi output current conveyor,” in TENCON 2019-2019 IEEE Region 10 Conference (TENCON), 2019, pp. 2179-2183. https://doi.org/10.1109/TENCON.2019.8929631. 26. S. Roy,T. K. Paul, S. Maiti, and R. R. Pal,“Voltage Dif­ferencing Current Conveyor Based Voltage-Mode and Current-Mode Universal Biquad Filters with Electronic Tuning Facility,” International Journal of Engineering and Technology Innovation, vol. 11, p. 146, 2021. https://doi.org/10.46604/ijeti.2021.6821. 27. M. Faseehuddin, J. Sampe, S. Shireen, and S. H. M. Ali, “Lossy and lossless inductance simulators and universal filters employing a new versatile active block,”Informacije MIDEM, vol. 48, pp. 97-114, 2018. 28. S. Singh, R. Tomar, and D. Chauhan, “A new cur­rent tunable current input current output biquad using CFTAs,” Journal of Engineering Science and Technology, vol. 12, pp. 2268-2282, 2017. 29. A. Kumar and S. K. Paul, “DX-MOCCII based fully cascadable second order current-mode universal fi lter,” Journal of Circuits, Systems and Computers, vol. 27, p. 1850113, 2018. https://doi.org/10.1142/S021812661850113X. 30. T. Ettaghzouti, N. Hassen, K. Garradhi, and K. Bes-bes, “SIMO high frequency active universal cur­rent mode filter with independent control of pole frequency and quality factor,” in 2017 18th Inter­national Conference on Sciences and Techniques of Automatic Control and Computer Engineering (STA), 2017, pp. 157-162. https://doi.org/10.1109/STA.2017.8314934. 31. S. Maheshwari, “Current conveyor all-pass sec­tions: brief review and novel solution,” The Scien­tific World Journal, vol. 2013, 2013. https://doi.org/10.1155/2013/429391. Copyright © 2022 by the Authors. This is an open access article dis­tributed under the Creative Com­mons Attribution (CC BY) License (https://creativecom-mons.org/licenses/by/4.0/), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Arrived: 19. 06. 2022 Accepted: 08. 11. 2022 Original scientific paper https://doi.org/10.33180/InfMIDEM2022.402 Electronic Components and Materials Vol. 52, No. 4(2022), 215 – 226 Influence of Trace Eelements on the Electrical Properties of ZnO-based Multilayer Varistors Slavko Bernik1, Nana Brguljan1, Marija Ercegovac2, Zoran Samardžija1 1Department for Nanostructured Materials, Jožef Stefan Institute, Ljubljana, Slovenia 2Bourns, d.o.o., Žužemberk, Slovenia Abstract: Nonlinear current-voltage (I-U) characteristics and stability after an IMAX test of two types of multilayer varistors (MLVs), each type fabricated in two series, were analysed in terms of their structure, microstructure and the presence of trace (i.e., impurity) elements. The structural and microstructural features showed nothing significant that could justify the very different IMAX characteristics of the MLVs of the same type from the two series. In the larger MLVs, declared for IMAX 1000A, the most critical factor was found to be the amount of Fe, the source of which was the starting Cr2O3 powder; one batch of Cr2O3 used for their fabrication contained an about 5-times-larger amount of Fe than the other, while the amounts of the other impurity elements (i.e., Al, Si, Mg, Ca, Ti, Na, K) were similar in both. The MLV1000 samples prepared with the Fe-rich Cr2O3 powder failed after a current impulse of 900A, while the samples using the Fe-low Cr2O3 powder withstood even 1400A. In the smaller MLVs, declared for 200A, prepared from Fe-low Cr2O3 and added in half the amount as in the MLV1000 samples, the critical factor was the large addition of SiO2 in the starting composition and the samples failed after a current impulse of 30 A. Amending the composition with the addition of several 100 ppm of Al resulted in an enhancement of IMAX to 420A, demonstrating the positive effects of Al. The results indicated the need to control the presence of trace elements and showed the complexity of an issue that requires a thorough consideration for each type of MLV to achieve the required electrical characteristics. Keywords: ZnO; multilayer varistors; trace elements; microstructure; electrical characteristics Vpliv elementov v sledovih na elektricne lastnosti vecplastnih varistorjev na osnovi ZnO Izvlecek: Nelinearne tokovno-napetostne (I-U) karakteristike in stabilnost po testu IMAX dveh vrst vecslojnih varistorjev (MLV), od katerih je bila vsaka izdelana v dveh serijah, smo analizirali glede na njihovo strukturo, mikrostrukturo in prisotnost elementov v sledovih (tj. necistoc). Strukturne in mikrostrukturne znacilnosti niso pokazale nic pomembnega, kar bi lahko pojasnilo zelo razlicne znacilnosti IMAX med MLV vzorci istega tipa iz obeh serij. Pri vecjih MLV, deklariranih za IMAX 1000A, se je izkazalo, da je najbolj kriticen dejavnik kolicina Fe, katerega vir je bil zacetni prah Cr2O3; prah Cr2O3 ene serije, uporabljen za njihovo izdelavo, je vseboval približno 5-krat vecjo kolicino Fe kot prah druge serije, medtem ko so bile kolicine ostalih necistoc (tj. Al, Si, Mg, Ca, Ti, Na, K) v obeh podobne. Vzorci MLV1000, izdelani s prahom Cr2O3, bogatim s Fe, so odpovedali že po tokovnem impulzu 900 A, medtem ko so vzorci, pripravljeni s prahom Cr2O3 z malo Fe, zdržali celo 1400 A. Pri manjših MLV, deklariranih za 200 A, pripravljenih iz Cr2O3 z nižjo vsebnostjo Fe, ki je bil dodan v polovicni kolicini kot pri vzorcih MLV1000, je bil kriticen dejavnik visok dodatek SiO2 v zacetni sestavi, tako da so vzorci odpovedali že po tokovnem impulzu 30 A. Sprememba sestave z dodatkom vec 100 ppm Al je povzrocila izboljšanje IMAX na 420A, kar dokazuje pozitivne ucinke Al. Rezultati so pokazali na pomembnost nadzora prisotnosti elementov v sledovih in na kompleksnost problematike, ki zahteva temeljit premislek za vsako vrsto MLV, da bi dosegli zahtevane elektricne karakteristike. Kljucne besede: ZnO; vecplastni varistorji; necistoce; mikrostruktura; elektricne lastnosti * Corresponding Author’s e-mail: slavko.bernik@ijs.si 1 Introduction broad range of operating voltages for electrical de-Varistors, i.e., variable resistors are core elements of vices and electronics, as well as for the stabilisation of surge protection devices (SPDs), complying with a low-, medium, and high-voltage electric power lines. How to cite: Slavko Bernik et al., “Influence of Trace Eelements on the Electrical Properties of ZnO-based Multilayer Varistors", Inf. Midem-J. Microelec-tron. Electron. Compon. Mater., Vol. 52, No. 4(2022), pp. 215–226 Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 The varistors are made of ZnO-based varistor ceram­ics, which are characterised by an exceptional current-voltage (I-U) nonlinearity and a high energy-absorp­tion capability. Accordingly, at the breakdown voltage the varistor switches from a highly resistive to a highly conductive state in a matter of nanoseconds and the current through the varistors increases by several or­ders of magnitude for a minimum change in voltage. Thus, a varistor connected in parallel effectively diverts the transient surge from the protected device to the ground and absorbs the excess harmful energy, ensur­ing the undisturbed and safe operation of the device, while preventing its damage or even destruction. The possibilities to tailor the break-down voltage of varistor ceramics from a few volts up to several kilo-volts ena­bles suitable dimensions of varistors for applications across a broad range of voltages, a superior response time to transient surges, a high energy-absorption capability and a long-term operating stability and reli­ability. These advantages of varistors mean that they are effectively unmatched by any other surge-protection device. Hence, varistors dominate the worldwide, multi-billion euro business of overvoltage-protection applica­tions [1-3]. The breakdown voltage of a varistor is proportional to the thickness of the ceramic and the energy-absorp­tion capability is proportional to its volume. Hence, so-called bulk or disc varistors are predominantly used for medium- and high-voltage applications, which are also related to higher energies. For breakdown voltages be­low about 60V, thicknesses below 1 mm are required and the fabrication of such thin ceramic discs without any shape deformation during high-temperature sin-tering can be difficult, and there is a problem with their low fracture strength. With an increasing thickness-to-diameter ratio of the discs to increase their volume, and hence the energy-absorption capability, the prob­lem of their low fracture strength further increases. The solution for low-voltage applications is multilayer varistors (MLV), which were developed in the 1980s as an answer to problems in low-voltage circuits, accom­panied by a rapid trend for their miniaturisation and a constant demand for an enhanced integration scale in electronic circuits. The miniaturisation of electronic cir­cuits increases their sensitivity to external interference and MLVs are used for low-voltage protection against transient surges caused by electrostatic discharge, at­mospheric discharge and transient overvoltages gen­erated for other reasons in integrated circuits, hybrid circuits and surface-mounted circuits [4-7]. Accord­ingly, MLVs also have key role in automotive-circuit protection as modern vehicles employ a variety of elec­tronics for safety, assisted driving, self-driving, camer­as, engine-performance optimisation with an engine-control unit, communications and navigation. Many of these systems require multiple processors as well as high-current sensors and actuators. Nowadays vehicles contain over 40 motors and actuators to drive win­dows, doors, seats, pumps, windshield wipers and oth­er components. At the standard operating voltages of personal cars and trucks, i.e., 12V and 24V, respectively, the automotive MLVs with nominal voltages from 20 to 40V are exposed to extremely heavy loads that occur for instance when turning on or off the engine, or when any other power user is switched on (i.e., electrical ad­justment of a seat, opening a window, etc.), which can cause transient voltage surges up to several 100 V last­ing for several 100 ms and energy loads of more than 100 J. However, in hybrid electric vehicles using 48 V systems and plug-in electric cars using high-voltage systems (i.e., 400V and even higher) because the high voltage boosts efficiency and allows lower current for the same power (wattage), the requirements for MLVs are even greater. It should also be mentioned that MLVs have to show no deterioration in performance in the temperature range from – 55°C to 175°C. Accordingly, a great deal of attention has to be given to any detail of the fabrication technology for MLVs to comply with the rigorous performance requirements [8]. MLVs or “chip” varistors are composed from layers of fine-grained ZnO-based varistor ceramics with thick­nesses typically in the range from 20 to 100 .m, inter­nal electrodes connected in parallel and terminal outer electrodes, and are manufactured using multilayer fab­rication technology. Flexible green ceramic tapes are prepared with tape-casting technology. Several green sheets with screen-printed inner electrodes (typically AgPd,) are stacked, isostatically laminated, diced into individual varistors, and co-sintered, typically at tem­peratures around 1000°C. The layers are laminated in a way that the inner electrodes form alternating con­nections between the two terminal electrodes; accord­ingly, every other layer connects to the same terminal electrode. Such a configuration allows for higher re­sistances at lower voltages with faster response times than the bulk metal oxide varistors (MOV). Basically, the breakdown or nominal voltage of MLVs is determined by the thickness of the ceramic layer between the two inner electrodes, while their energy-absorption capa­bility can be adjusted with the number of layers (i.e., the thickness of MLV) and the area of MLV, i.e., by ad­justing its volume [5, 9]. Among the generally well-known types of the ZnO-based varistor ceramics, i.e., ZnO-Bi2O3-based, ZnO-PrO-bazed, and ZnO-VO-based, the ZnO-BiO­ 6112523 based ones are the most widely used for bulk varistors and MLVs. ZnO is an n-type, wide-band-gap semicon­ductor. The current-voltage (I-U) nonlinearity, which is typically described with the expression (0) Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 (0) IkU a  (k – constant, . – coefficient of nonlinearity), is induced to ZnO by the addition of varistor formers like Bi2O3; it segregates at the ZnO grain boundaries and results in the formation of electrostatic Schottky barriers so that the non-ohmic (i.e., varistor) grain boundary has a breakdown voltage UGB of about 3.2V. Varistor form­ers facilitate the formation of acceptor states at the grain boundaries, i.e., oxygen interstitials (Oi’’) and zinc vacancies (VZn’, VZn’’), which act as electron traps, while in the vicinity of grain boundaries, in the ZnO grain, a positively charged depletion layer of oxygen vacancies (V., V..) and zinc interstitials (Zn., Zn..) is formed. The OO ii weak nonlinearity of Bi2O3-doped ZnO with . of about 3 is further enhanced by the addition of dopants like SbO, CoO, MnO, NiO and CrO to values of . from 23343423 20 to 80. While some of these dopants are essential to increase the electrical conductivity of the ZnO grains (Co, Mn, Ni), the other (Sb, Cr) affect the growth of the ZnO grains and thus enable tailoring of the breakdown voltage (UB) of varistor ceramics via grain size (G) in ac­cordance with the expression (1) UGB * t U U * N (1) B GB GB G where t is thickness of ceramics and NGB number of grain boundaries. Although the varistor dopants added to ZnO typically account in total for less than 10 wt.%, such a composition results in a rather complex micro-structure of the varistor ceramics. It contains the ZnO phase and secondary phases of the ZnO-Bi2O3-Sb2O3 system, i.e., a BiO-rich phase, a ZnSbO-type spinel 237212 phase and a BiZnSbO-type pyrochlore phase, while 32314 the other varistor dopants (i.e., Co, Mn, Ni, Cr) are incor­porated into these phases. The electrical characteristics of the varistor ceramics, i.e., breakdown voltage (UB), coefficient of nonlinearity (.) and leakage current (IL), are primarily affected by the ZnO phase, which must be highly conductive, and the Bi2O3-rich phase at the grain boundaries for their non-ohmic varistor charac­teristic (highly resistive at voltages below UB). It is also important that the Bi2O3 forms a liquid phase during sintering and, besides the ZnO, dissolves all the other varistor dopants, thus greatly affecting their distribu­tion in the microstructure as well as the sintering and the grain-growth process. Accordingly, all the dopants in equilibrium amounts are incorporated into the Bi2O3­rich phase, thus affecting the electronic states at the grain boundaries and consequently also their electrical characteristics [1, 2, 6, 7]. Some elements can greatly affect the electrical charac­teristics of the varistor ceramics in very small amounts of up to only several 100ppm. They can act as donors, acceptors, or both, depending on the nature, concen­tration and location in the host crystal lattice. In the microstructure they can be grain boundary or grain specific and accordingly they affect the current-voltage (I-U) characteristics in the “pre-breakdown” region at low currents or in the “upturn” region at high currents, or both. The controlled addition of such carefully se­lected elements can be used for a targeted improve­ment of the electrical and energy characteristics of varistor ceramics. For example, fine doping with Al is generally used for the enhanced stability of varistor ce­ramics at high currents due to the improved electrical conductivity of the ZnO grains; however, it also increas­es the leakage current. In contrast, the fine addition of Si increases the resistivity of the grain boundaries, thus it is often used to decrease the leakage current of varis­tor ceramics. Unfortunately, elements that strongly af­fect the functional properties of varistor ceramics, even in very small quantities, can also be added unintention­ally as impurities of the standard varistor dopants, which represents a serious problem. Hence, it is important to control the presence of impurity elements in the oxide powders used for the fabrication of varistor ceramics [10-12]. In this work two types of the multilayer varistors (MLV) from two fabrication series were analysed and their current-voltage (I-U) and energy characteristics are dis­cussed in terms of their structure, microstructure and the presence of impurity (i.e., trace) elements, the pri­mary source of which was confirmed to be the start­ing Cr2O3 powder. In the larger type of MLVs, used for maximum current impulses of IMAX=1000A, the most critical for the stability against high current impulses was the amount of Fe impurity present. At a much low­er amount of Fe (< 10 ppm) and a similar amount of other impurities (i.e., Al, Si, Mg, Ca, Ti, Na, K) the charac­teristics of this type of MLV were excellent with an IMAX of 1400A, while MLVs prepared from Fe-rich Cr2O3 and thus containing about 40 ppm of Fe failed at current impulses of 900A. In the case of smaller MLVs, declared for IMAX=200A, too much SiO2 added in the starting composition was found to be critical and they failed at 30A. However, amending their starting composi­tion with the addition of Al resulted in significantly im­proved energy characteristics, raising their IMAX to 420A. The results indicated the importance of controlling the presence of trace elements, which can critically affect the performance of MLVs. 2 Materials and methods Two types of MLVs, i.e., the larger, declared for a maxi­mum impulse current (IMAX) of 1000A (labelled MLV1000) and the smaller, declared for an IMAX of 200A (labelled Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 MLV200), were prepared with standard multilayer fab­rication technology in the Bourns company [4, 13]. In all the MLVs the same reagent grade powders of oxides of Zn, Bi, Sb, Co, Mn, Cr and Si were used; however, in the case of Cr2O3, powders from the same producer but from two different batches were used, both having the same composition according to the supplier, and are here labelled Cr1 and Cr2. For the MLV1000, ceramic tapes with the composition 97.9ZnO + 2.1(Bi2O3, Sb2O3, CoO, MnO, CrO) were used. One series of this type 343423 was prepared from tapes with Cr1 (series MLV1000­Cr1) and the other series from ceramic tapes with Cr2 (series MLV1000-Cr2). The smaller MLVs were prepared from ceramic tapes with the composition 98.0ZnO + 2.0(BiO, SbO, CoO, MnO, CrO, SiO); in one se­ 23233434232 ries the tapes containing Cr2 (series MLV200-Cr2) and in the other series the ceramic tapes also contained Cr2 but with their composition altered with addition of several 100 ppm of Al as solution of Al(NO3)3·9H2O (series MLV200-Cr2Al). It should be mentioned that in the MLV1000 the amount of added Cr2O3 was twice the amount added in the MLV200. In the MLV200 samples, however, also a large amount of about 0.1mol.% of SiO2 was added to the starting composition of the ceramic tapes, which was not added to the MLV1000. All the series of MLVs were co-sintered with AgPd electrodes in air at about 1000°C for the same duration, the series MLV200 at about 20 to 30°C lower temperature than the series of MLV1000 samples. The current-voltage (I-U) characteristics of the MLV sam­ples were measured using a Keithley 2410 Source Meter, i.e., nominal voltage (i.e., breakdown voltage), UN, was determined at a current of 1 mA, leakage current, IL, at 0.75UN, and the coefficient of nonlinearity, ., was deter­mined in accordance with the equation (2) I log 2  I  1  a  (2) U 2 l og  U  1  where U2 and U1 are voltages measured at I1 = 1 mA and I2 = 10 mA, respectively. The high current stability was determined by the maximum impulse current (IMAX) at which the UN of the MLVs changes by less than 10%. The IMAX of the samples was analysed using an AMC MIG0606 impulse generator with current impulses of shape 8/20 (i.e., rising time 8 .s, duration 20 .s) to simulate impulses caused by a lightning strike) at cur­rent intensities from 200A to 500A for series MLV200 and from 900A to 1500A for series MLV1000. The I-U characteristics of the samples were measured before and after the current impulse test and their average values determined based on measurements of at least 10 samples per test. Microstructures in the cross-section of the MLVs, per­pendicular to the plane of the internal electrodes, were prepared and analysed in the scanning electron micro­scope (SEM) JEOL JSM-7600F equipped with an ener­gy-dispersive spectrometer (EDS) Oxford Instruments INCA. The integrity of the inner electrodes, the distance between them and their connectivity to the terminal electrodes was examined. Also, the microstructure of the varistor ceramics was analysed with regard to the phase composition, phase homogeneity of the micro-structure, porosity, grain size and grain size distribution. The phase composition of the ceramics was also deter­mined with a powder x-ray diffraction analysis (XRD). Furthermore, starting oxide powders used for the preparation of the foils of varistor ceramic using tape-casting technology were examined for their crystal structure (XRD), grain size and grain size distribution (particle size analyser HORIBA), morphology of the particles (SEM analysis), and the chemical composition (SEM/EDS). Finally, for all the used oxide powders, a pre­cise quantitative analysis of the trace impurity elements was also made using the inductively coupled plasma ­optical emission spectrometry method (ICP-OES). 3 Results and discussion The average current-voltage (I-U) characteristics of the MLV1000 samples before and after the IMAX current im­pulse test are presented in Tables 1 and 2. For this type of MLV an IMAX of at least 1000A is required, and as can be seen in Table 1 none of the MLV samples from the series MLV1000-Cr1 complied with this requirement. In the first line of the table (i.e., shaded grey) the aver­age I-U characteristics of all the MLV samples from this series measured before the IMAX test are given. In com­parison to these reference values, already after current impulse of 900A the I-U characteristics deteriorated, as indicated by a decrease of the nominal voltage (UN) and the coefficient of nonlinearity (.), while the leak­age current (IL) significantly increased. The degradation of the I-U characteristics of the MLVs from this series further increased with a rising of the current impulse’s intensity so that after an impulse of 1200A the average UN decreased by 34 %, the . decreased by 50% from 34 to 17, and the IL increased strongly. In contrast, the samples from the series MLV1000-Cr2 (Table 2) showed excellent stability even after current impulses of 1300A and 1400A, and failed only after tests at 1500A. Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 Table 1: Current-voltage characteristics (I-U) of MLV1000 samples from series MLV1000-Cr1 before and after IMAX test. IMAX /A UN/V (±s/%) a (±s/%) IL/µA (±s/%) .UN /% Ref. 35.1 (3) 34 (6) 1.6 (70) / 900 29.1 (54) 25 (57) 201 (223) -17 1000 30.5 (38) 22 (61) 202 (221) -13 1100 24.3 (61) 16 (89) 403 (135) -30 1200 23.0 (74) 17 (88) 438 (119) -34 Such drastically different IMAX characteristics between the MLV1000 samples from two series could result from some failure in their fabrication, which would show in their microstructure. Accordingly, their microstructures were examined for possible defects to the internal elec­trodes, like not being whole or continuous, but with interruptions, uniformity of the distance between in­ternal electrodes, number of internal electrodes, poor or failed connection of the internal electrodes with ter­minal electrodes, and also in regard to the microstruc­ture and phase composition of the varistor ceramics. The microstructures of several MLV1000 samples from each series were examined on the SEM and nothing significant that could explain the different electrical characteristics was found. Typical microstructures of the MLV1000 samples in the cross-section direction are presented in Fig. 1. Table 2: Current-voltage characteristics (I-U) of MLV1000 samples from series MLV1000-Cr2 before and after IMAX test. IMAX /A UN/V (±s/%) a (±s/%) IL/µA (±s/%) .UN /% Ref. 33.0 (2) 33 (4) 1.4 (36) / 1300 32.9 (4) 32 (4) 2.9 (70) 1 1400 33.4 (1) 32 (3) 3.9 (54) 1 1500 24.6 (58) 20 (81) 402 (136) -27 The internal electrodes, 7 in total, were found in all the analysed samples from both series to be continu­ous, at a uniform distance of about 104 .m, and all well in contact with the terminal electrodes. Also, the SEM analysis of the varistor ceramics showed no dif­ference in the microstructure of the MLV1000 samples from both series in terms of porosity, phase composi­tion and homogeneity in the distribution of second­ary phases among the ZnO grains. In all the samples, besides the matrix ZnO phase, also a secondary Bi2O3­rich liquid phase, a Zn7Sb2O12-type spinel phase, and a BiZnSbO-type pyrochlore phase were determined 32314 by the EDS analysis. In Fig. 2, typical microstructures and phase compositions of the samples MLV1000-Cr1 (2.a-b) and MLV1000-Cr2 (1.c-d) are shown. SEM analy­sis of the etched microstructures (Fig. 3) showed that the samples MLV1000 from both series also have simi­lar ZnO grain sizes and size distributions (Fig. 3). Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 Figure 3: SEM/BE images of the etched microstructures of the MLV samples; (a,b) MLV1000-Cr1, (c,d) MLV1000­Cr2, and (e,f) MLV200-Cr2. The XRD analysis also confirmed the same phase com­position of the MLV1000 samples from both series, showing besides the ZnO phase also a Bi2O3-rich liquid phase as a .-BiO modification and a ZnSbO-type 237212 spinel phase. The XRD peaks of the BiZnSbO-type 32314 pyrochlore phase, which was found by the SEM/EDS analysis in the microstructures of the MLV1000 sam­ples, strongly overlap with the other phases present. Hence, it is difficult to detect pyrochlore phase in the varistor ceramics by the XRD analysis. The typical XRD pattern of the varistor ceramics in the analysed MLV samples is shown in Fig. 4. The average I-U characteristics of the smaller MLV200 samples before (i.e., reference values) and after the cur­rent impulse test are given in Tables 3 and 4. The sam­ples of the series MLV200-Cr2 showed extremely poor IMAX characteristics (Table 3); while they should with­stand a current impulse of 200A with a change in the UN of less than 10% at a preserved coefficient of nonlin­earity (.) and a low leakage current (IL), even a current impulse of just 30A resulted in a significant decrease of UN by 18%, accompanied with a decrease in . and a significant increase in IL. Table 3: Current-voltage characteristics (I-U) of the MLV200 samples from the series MLV200-Cr2 before (reference values) and after IMAX test. IMAX /A UN/V (±s/%) a (±s/%) IL/µA (±s/%) .UN /% Ref. 28.1 (3) 29 (6) 0.2 (106) / 30 23.0 (53) 24 (53) 200 (224) -18 50 23.4 (51) 24 (53) 200 (224) -17 100 12.9 (61) 13 (121) 600 (91) -30 The samples from the series MLV200-Cr2Al (Table 4), however, showed excellent stability for current impuls­es, even up to 420A, while after a load with 440 A their I-U characteristics significantly decreased and even more after a current impulse of 500A. Table 4: Current-voltage characteristics (I-U) of the MLV200 samples from the series MLV200-Cr2Al before (reference values) and after IMAX test. IMAX /A UN/V (±s/%) a (±s/%) IL/µA (±s/%) .UN /% Ref. 26.9 (4) 32 (8) 1.5 (109) / 200 28.1 (2) 33 (10) 0.6 (24) 1 260 28.7 (2) 31 (9) 0.4 (28) 1 300 27.1 (4) 31 (8) 1.5 (121) 1 340 27.1 (3) 31 (19) 1.4 (11) 1 380 26.5 (3) 26 (14) 1.7 (38) 2 400 26.7 (2) 28 (31) 1.9 (68) 2 420 26.8 (1) 30 (4) 2.3 (31) 1 440 19.4 (56) 18 (80) 402 (136) -27 500 11.5 (119) 10 (121) 647 (76) -56 Microstructural analysis of the MLV200 samples from both series showed that all six internal electrodes have a similar thickness and are continuous, even at a dis­tance of about 80 .m, and well connected to the ter­minal electrodes (Fig. 5). Also, the SEM/EDS analysis of the varistor ceramics showed similar microstructures in terms of the phase composition and the grain size (Figs. 2.e-f, and 3.e-f). The same phase composition of Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 the varistor ceramics in the MLV200 samples from both series was also confirmed by the XRD analysis (Fig. 4). Figure 5: SEM/BE images showing typical cross-section microstructure of the samples MLV200. The results showed that in both types of MLVs, the MLV1000 and MLV200 of both series, varistor ceramics have similar microstructures and phase compositions according to the SEM/EDS and XRD analysis. Actually, these analyses revealed nothing related to the micro-structure of the ceramics and the structure of the MLVs, including possible technical errors in their fabrication, that could explain such drastically different stabilities in the I-U characteristics after the current impulse tests (IMAX test) between the same type of MLVs from two fabrication series. However, such results indicated that attention should be given to know the differences in the preparation of the MLV samples, i.e., in the case of the MLV1000 samples the use of Cr2O3 from different batches and in the case of MLV200 samples an altera­tion of the composition with the addition of Al in one fabrication series as compared to the other, while in both the Fe-low Cr2O3 from the same batch was used. Accordingly, the Cr2O3 powders used for the prepara­tion of the MLV samples were thoroughly analysed. Granulometric analyses of the Cr2O3 powders from both batches showed that they have similar average particle sizes of about 2.1 .m and also similar particle size distributions in the range from 0.2 .m to 10 .m, as shown in Fig. 6. The typical morphology of the used Cr2O3 powders is shown in Fig. 7 and compliments the results of the granulometric analysis. Figure 7: SEM images showing typical morphology of the Cr2O3 powders used for the fabrication of the MLV samples. The XRD patterns of both powders are very similar (Fig. 8) and can be identified using the reference pattern for Cr2O3 JCPDF 00-038-1479. However, in the Cr1 powder additional minor peaks indicate the presence of some SiO2 secondary phase (JCPDF01-082-1556). Detailed SEM/EDS analyses revealed, in both Cr2O3 pow­ders, the presence of a significant amount of secondary phases as coarse-grained inclusions containing impurity elements, primarily Si, Al and Fe, and also K, Na, Ca and Ti (Fig. 9). Hence, a quantitative chemical ICP-OES analysis was made to determine the content of trace impurity el­ements in the Cr2O3 powders. The results of the ICP-OES analysis of the Cr2O3 from both batches are presented in Table 5. Figure 8: XRD patterns of the Cr2O3 powders; in the Cr1, additional to the peaks of Cr2O3, also additional mi­nor peaks indicate the presence of the SiO2 secondary phase. The quantitative ICP-OES analysis of all the other oxides used for the fabrication of the studied MLV samples (i.e., ZnO, SbO, CoO, and MnO) showed the presence of 233434 the impurity elements below 50 ppm, confirming that the main source of impurities is CrO 23. Figure 6: Histogram of the particle size distribution in the Cr2O3 powder. Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 Table 5: Results of the quantitative ICP-OES analysis of the used Cr2O3 powders, i.e., Cr1 and Cr2, for the amount of impurity elements. Composition Cr2O3 powder M(ppm)/MO(wt.%) Cr1 Cr2 Cr2O3 97.08 97.99 Si/SiO2 10080/1.57 10020/1.49 Al/Al2O3 3502/0.12 2801/0.10 Fe/Fe2O3 9488/0.92 1944/0.19 Mg/MgO 932/0.10 779/0.09 Ca/CaO 888/0.08 797/0.08 Ti/TiO2 172/0.02 127/0.01 Na/Na2O 477/0.04 231/0.02 K/K2O 870/0.07 351/0.03 These results showed the contamination of both Cr2O3 powders used in the preparation of the MLVs, with numerous impurity elements that are known to have an influence on the current-voltage (I-U) characteris­tics and the stability of the varistor ceramics. Several models are proposed, explaining the voltage stability/ instability of the ZnO-based varistor ceramic through the degradation of the electrostatic barriers at the grain boundaries. However, it is common to most of them that the origin of the degradation is assigned to the dif­fusion of zinc interstitials (Zni) from the depletion layer and their chemical interaction with acceptor states at the grain boundaries, i.e., oxygen interstitials (Oi) and zinc vacancies (VZn), which consequently leads to the degradation and collapse of the electrostatic Schottky barriers [1, 10, 11, 14-16]. Their degradation is typically expressed by a decrease in the nominal (i.e., break­down) voltage of the varistor ceramics (UN) due to the reduced breakdown voltage of the grain boundaries (UGB), a decrease in the coefficient of nonlinearity (.) and a significant increase of the leakage current (IL), as observed in the case of the poor MLVs in this work. Ac­tually, most of the impurity elements detected in the Cr2O3 are reported in the literature as having a positive influence in low amounts and some of them are even known as “varistor highlighters”, like Al and Si, which are intentionally added to the starting composition in ppm amounts to enhance the performance of the varistor ceramics in the low-current pre-breakdown region or in the high-current “up-turn” region of their I-U curve [2]. They affect the defect equilibria and electronic states at the grain boundaries, and thus the height and stability of the electrostatic Schottky barri­ers at the grain boundaries, which are responsible for the I-U nonlinearity of varistor ceramics. On the other hand, they can increase the electrical conductivity of the ZnO grains, which has a positive influence on the stability and aging characteristics of varistor ceramics due to lowering the released Ohmic heat under high current loads [1, 10]. The main impurity detected in Cr2O3 powders is cer­tainly Si (i.e. SiO2), as can be seen in Table 5. Studies showed that SiO2 in low amounts strongly affects the electronic states at the grain boundaries, resulting in an increase of the height of the electronic Schottky barriers and also the coefficient of nonlinearity (.). Also, the increased resistivity of the grain boundaries results in a lower leakage current (IL). At the same time, SiO2 also increases the depletion-layer width, leading to the enhancement of the breakdown voltage of the grain boundaries (UGB). However, too much Cr2O3 could result in a deterioration of the I-U characteristics, espe­cially once the secondary Zn2SiO4 phase starts to form at the grain boundaries [12, 17, 18]. Another impurity element detected in the Cr2O3 powders in amounts of several 1000ppm is Al; it is often considered at the main “varistor highlighter” and also generally known as the main donor dopant in ZnO-based ceramics. Numer­ous studies showed that it enhances the energy stabil­ity of the ZnO-based varistor ceramics when added in amounts of several 100 ppm, by increasing the con­ductivity of the ZnO grains. However, Al is not a grain selective dopant, but also influences the electronic states at the grain boundaries, resulting in an increase of the leakage current (IL). Also, at higher amounts of Al, which depends on the sintering temperature and time, it starts to incorporate at the interstitial sites in the crystal structure of ZnO, acting as an acceptor and decreasing the conductivity of the ZnO grains. Hence, thorough control of the amount of Al is required, de­pending on the sintering conditions [11, 19-22]. The amphoteric dopants in ZnO are Na and K. In very low amounts of a few 10 ppm, Na (and K) incorporate at the Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 interstitial sites of the ZnO crystal lattice and act as do­nors, increasing the electrical conductivity of the ZnO grains, while at the same time they substitute for the interstitial Zn (Zni) in the depletion layer, decreasing its concentration and thus increasing the stability of ZnO varistor ceramics. However, in larger amounts, Na (K) incorporates at the regular sites of Zn in the structure of ZnO (NaZn’) and acts as an acceptor, decreasing the electrical conductivity of the ZnO grains [10, 23, 24]. The positive influence of low levels of doping was also reported for Mg [24, 25] and Ca [24, 26-28]; both of them result in a decrease of the leakage current (IL), an increase of the breakdown voltage (UB) and an increase of the coefficient of nonlinearity (.). In the case of Ca it has been reported that it increases the solid solubility of Al in the ZnO, thus enhancing the electrical conduc­tivity of the ZnO grains, while reducing the accumula­tion of Al at the grain boundaries, which increases their resistivity [28]. If added in too large amounts, each of them results in secondary phases at the grain bounda­ries of the ZnO, causing deterioration of the I-U char­acteristics of varistor ceramics; in the case of Mg the Mg-Zn-O periclase solid solution is formed [25], and in the case of Ca the Ca-Bi-O phases [26]. In contrast to other impurity elements found in Cr2O3, which have a positive influence, Fe has a negative influence on the I-U characteristics of the varistor ceramics already in low amounts [29-32], decreasing the breakdown volt­age (UB) and the coefficient of nonlinearity (.), and increasing the leakage current (IL). Peiteado et al. [30] explained such influence of Fe by its incorporation into the Zn7Sb2O12-type spinel phase at the grain bounda­ries, which strongly increases the electrical conductiv­ity of otherwise insulating secondary phase, and thus increases the conductivity of varistor ceramics below the breakdown voltage, enhancing their electrical deg­radation. Some of the impurity elements are present in the Cr2O3 powders in significant amounts of several 1000 ppm, and if present in such amounts in the varistor ceram­ics, they would likely have a negative influence on the I-U characteristics. However, for the amounts of Cr2O3 in the composition of the varistor ceramics, 1000 ppm of the impurity element in Cr2O3 means about 4 ppm in the varistor ceramics. Accordingly, most of the impurity elements, which are introduced by Cr2O3 into the varis­tor ceramics of the studied MLV samples, are present in very low amounts. This can have a positive influence on their I-U characteristics, if any. Only Fe is known to have a negative influence and in the I-U characteristics of varistor ceramics, even in very small amounts. In the samples from the series MLV1000-Cr1, the Fe is present in the amount of almost 40 ppm and in the same type MLV samples from the series MLV1000-Cr2 only in the amount of about 8 ppm. Accordingly, too much Fe in the MLV1000-Cr1 samples is the likely reason for their poor IMAX characteristic, when they failed at a current impulse of 900A (Table 1). In the samples MLV1000-Cr2, however, the amount of Fe is below some critical value; therefore, they can have an excellent IMAX of 1400A (Table 2). In the smaller MLV200 samples, the amount of add­ed Cr2O3 is half the amount added in the samples MLV1000-Cr2. Hence, the amount of impurity elements in the varistor ceramics of MLV200 samples is even low­er, only about 2 ppm for 1000 ppm present in Cr2O3. Accordingly, impurity elements probably have even less influence on the I-U characteristics of the MLV200 samples than on the MLV1000-Cr2 samples, but in the case of most of them, it is likely a positive one. This in­dicates that the poor IMAX characteristics of the sample from the MLV200-Cr2 series (Table 3) is likely caused by too much SiO2 in the starting composition of the varis­tor ceramics. However, the much better performance of the samples from the series MLV200-Cr2Al (Table 4), having the starting composition of the varistor ce­ramics corrected with the addition of Al in the optimal amount of several 100 ppm, indicates the positive ef­fect of Al, compensating the negative effect of the ex­cess Si, and thus increasing the IMAX to even 420A. 4 Conclusions Two types of ZnO-Bi2O3-based multilayer varistors (MLVs), declared for different maximum current im­pulses (IMAX), were studied. Their current-voltage (I-U) and energy characteristics (IMAX test with current im­pulses 8/20) were analysed in terms of their structure, microstructure, starting composition and presence of the impurity (i.e., trace) elements. The larger MLV samples, declared for the IMAX 1000A (MLV1000), which were fabricated in two series using Cr2O3 from different batches, showed dramatically dif­ferent current-voltage (I-U) characteristics after IMAX tests. The samples MLV1000 from one series failed already after a current impulse of 900A, while the samples from the other series preserved the I-U char­acteristics even after a load with a current impulse of 1400A. Detailed analysis of their microstructure, phase composition and internal structure showed nothing significant that could explain such large differences in IMAX between the MLV1000 samples from both series. The compositional analysis showed that the used Cr2O3 powders contained a number of impurity elements in amounts from several 1000 ppm (Al, Fe) to even 10,000 ppm (Si), and from a few 100 ppm to 1000 ppm (Mg, Ca, Ti, Na, K). While the amount of most impurity elements was similar in the Cr2O3 powders from both series, Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 the amount of Fe in one powder was almost 5 times higher (9500 ppm) than in the other (1950 ppm). For the amount of Cr2O3 in the composition of varistor ce­ramics, 1000 ppm of the impurity element in the Cr2O3 means about 4 ppm in the varistor ceramics. Most of these elements are known to enhance the nonlinear I-U characteristics and the stability of the ZnO-Bi2O3­based varistor ceramics when present in such low amounts as introduced by the Cr2O3. The exception is Fe, which is known to degrade the I-U characteristics already for very small amounts. The difference in the IMAX of the MLV1000 samples from the two series can be attributed to different content of Fe. In the poor series of MLV1000 samples, the Fe-rich Cr2O3 powder resulted in almost 40 ppm of Fe in the varistor ceramics and consequently a degradation of their I-U characteristics even for current impulses below the declared 1000A. For comparison, in the good MLV1000 series, the Fe-low Cr2O3 powder introduced less than 10 ppm of Fe to the varistor ceramics. In the case of smaller MLVs, declared for 200A, with the Fe-low Cr2O3 added in half the amount as in the samples MLV1000 and with addition of SiO2, also significant dif­ferences in IMAX characteristics were observed between the two fabrication series, one with and one without the addition of Al. The samples from the series without added Al failed already after a current impulse of 30 A. In contrast, the samples from the other series having a starting composition of the varistor ceramics amend­ed with the addition of several 100ppm of Al endured current impulses with the shape 8/20 even up to 420A without changes in their I-U characteristics and failed only at higher current impulses. Such results indicated that the addition of Al neutralized the negative effect of the too large amount of added SiO2 and significantly improved the IMAX characteristics of the MLV200 sam­ples. The results show the importance of controlling the presence of trace elements, the source of which can be the starting raw materials for the preparation of varistor ceramics, or they could be intentionally added in the starting composition, as well as understanding their influence so as to achieve the required MLV properties. 5 Acknowledgments This work was supported by the European Regional Development Fund and Ministry of Education, Sci­ence and Sport of Republic of Slovenia (Project Grants C3330-18-952024). 6 Conflict of Interest The authors declare no conflict of interest in connec­tion to the work presented. 7 References 1. D. R. Clarke, “Varistor ceramics,” Journal of the American Ceramic Society, vol. 82, no. 3, pp. 485 - 502, 1999. 2. T. K. Gupta, “Application of zinc oxide varistors,” Journal of the American Ceramic Society, vol. 73, no. 7, pp. 1817-1840, 1990. 3. “Global Metal Oxide Varistor Market Size - Global Industry Size, Share, Trends, Competitions and Forecast, 2017-2022,” 2022.doi: https://www.tech­sciresearch.com/report/metal-oxide-varistor­market/3674.html 4. D. Szwagierczak, J. Kulawik, and A. Skwarek,“Infl u-ence of processing on microstructure and electri­cal characteristics of multilayer varistors,” Journal of Advanced Ceramics, vol. 8, no. 3, pp. 408-417, 2019. https://doi.org/10.1007/s40145-019-0323-7 5. L. Wang, G. Tang, and Z.-K. Xu, “Preparation and electrical properties of multilayer ZnO varistors with water-based tape casting,” Ceramics Interna­tional, vol. 35, no. 1, pp. 487-492, 2009. https://doi.org/10.1016/j.ceramint.2008.01.011 6. W.-H. Lee, W.-T. Chen, Y.-C. Lee, S.-P. Lin, and T. Yang, “Relationship between Microstructure and Electrical Properties of ZnO-based Multilayer Var­istor,” Japanese Journal of Applied Physics, vol. 45, no. 6A, pp. 5126-5131, 2006. https://doi.org/10.1143/jjap.45.5126 7. S. Hirose, K. Nishita, and H. Niimi, “Influence of dis­tribution of additives on electrical potential barri­er at grain boundaries in ZnO-based multilayered chip varistor,” Journal of Applied Physics, vol. 100, no. 8, 2006. https://doi.org/10.1063/1.2358833 8. (2019). Multilayer varistors in automotive circuit protection. Available: https://electronics360. globalspec.com/article/14398/multilayer-varis­tors-in-automotive-circuit-protection 9. J. Kulawik, D. Szwagierczak, and A. Skwarek, “Elec­trical and microstructural characterization of doped ZnO based multilayer varistors,” Microelec­tronics International, vol. 34, no. 3, pp. 116-120, 2017. https://doi.org/10.1108/mi-02-2017-0009 10. T. K. Gupta and A. C. Miller, “Improved stability of the ZnO varistor via donor and acceptor doping Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 at the grain boiundary,” Journal of Material Re­search, vol. 3, no. 4, pp. 745-751, 1988. 11. T. K. Gupta, “Microstructural engineering through donor and acceptor doping in the grain and grain boundary of a polycristalline semiconducting c eramics,” Journal of Material Research, vol. 7, no. 12, pp. 3280-3295, 1992. 12. B. Wang, Z. Fang, Z. Fu, and Y. Peng, “Electrical Be­havior against Current Impulse in ZnO Varistor Ceramics with SiO2 Addition,” Journal of Physics: Conference Series, vol. 1637, no. 1, 2020. https://doi.org/10.1088/1742-6596/1637/1/012026 13. J. Sun, B. Luo, and H. Li, “A Review on the Conven­tional Capacitors, Supercapacitors, and Emerging Hybrid Ion Capacitors: Past, Present, and Future,” Advanced Energy and Sustainability Research, vol. 3, no. 6, 2022. https://doi.org/10.1002/aesr.202100191 14. J. He, C. Cheng, and J. Hu, “Electrical degradation of double-Schottky barrier in ZnO varistors,” AIP Advances, vol. 6, no. 3, 2016. https://doi.org/10.1063/1.4944485 15. T. K. Gupta and W. G. Carlson, “A grain-boundary defect model for stability/instability of a ZnO varistor,” Journal of Materials Science, vol. 20, pp. 3487-3500, 1985. 16. Y. Huang, M. Guo, and J. Li, “Multiscale defect re­sponses in understanding degradation in zinc ox­ide varistor ceramics,” Ceramics International, vol. 46, no. 14, pp. 22134-22139, 2020. https://doi.org/10.1016/j.ceramint.2020.05.286 17. H. Bai et al., “Influence of SiO2 on electrical proper­ties of the highly nonlinear ZnO-Bi2O3-MnO2 varis-tors,” Journal of the European Ceramic Society, vol. 37, no. 13, pp. 3965-3971, 2017. https://doi.org/10.1016/j.jeurceramsoc.2017.05.014 18. Z. H. Wu, J. H. Fang, D. Xu, Q. D. Zhong, and L.-y. Shi, “Effect of SiO2 addition on the microstructure and electrical properties of ZnO-based varistors,” International Journal of Minerals, Metallurgy, and Materials, vol. 17, no. 1, pp. 86-91, 2010. https://doi.org/10.1007/s12613-010-0115-0 19. W. Long, J. Hu, J. Liu, J. He, and R. Zong, “The Ef­fect of Aluminum on Electrical Properties of ZnO Varistors,” Journal of the American Ceramic Society, vol. 93, no. 9, pp. 2441-2444, 2010. https://doi.org/10.1111/j.1551-2916.2010.03787.x 20. Q. Fu, C. Ke, Y. Hu, Z. Zheng, T. Chen, and Y. Xu, “Al-doped ZnO varistors prepared by a two-step dop­ing process,” Advances in Applied Ceramics, vol. 117, no. 4, pp. 237-242, 2018. https://doi.org/10.1080/17436753.2017.1405556 21. M. Houabes, S. Bernik, C. Talhi, and A. Bui, “The ef­fect of aluminium oxide on the residual voltage of ZnO varistors,” Ceramics International, vol. 31, no. 6, pp. 783-789, 2005. https://doi.org/10.1016/j.ceramint.2004.09.004 22 S. Bernik and N. Daneu, “Characteristics of .ZnO-based varistor ceramics doped with Al2O3,” Jour­nal of the European Ceramic Society, vol. 27, no. 10, pp. 3161-3170, 2007. https://doi.org/10.1016/j.jeurceramsoc.2007.02.176 23. M. Peiteado, Y. Iglesias, and A. C. Caballero, “So­dium impurities in ZnO–Bi2O3–Sb2O3 based varis-tors,” Ceramics International, vol. 37, no. 3, pp. 819­824, 2011. https://doi.org/10.1016/j.ceramint.2010.10.016 24. E. L. Tikhomirova, O. G. Gromov, and Y. A. Savel’ev, “Effect of Impurities on Varistor Properties of High-Voltage ZnO Ceramics,” Russian Journal of Applied Chemistry, vol. 94, no. 4, pp. 437-441, 2021. https://doi.org/10.1134/s1070427221040029 25. A. Smith, G. Gasgnier, and P. Abelard, “Voltage-Current Characteristics of ZnO varistors of a Sim­ple Zinc Oxide Varistor Containing Magnesia,” Journal of the American Ceramic Society, vol. 73, no. 4, pp. 1098-1099, 1990. 26. K. Hembram, T. N. Rao, M. Ramakrishana, R. S. Srini­vasa, and A. R. Kulkarni, “Influence of CaO doping on phase, microstructure, electrical and dielectric properties of ZnO varistors,” Journal of Alloys and Compounds, vol. 817, 2020. https://doi.org/10.1016/j.jallcom.2019.152700 27. H. Wang, H. Zhao, W. Liang, S. Fan, and J. Kang, “Ef­fect of sintering process on the electrical proper­ties and microstructure of Ca-doped ZnO varistor ceramics,” Materials Science in Semiconductor Pro­cessing, vol. 133, 2021. https://doi.org/10.1016/j.mssp.2021.105880 28. H. Zhao, H. Wang, X. Meng, J. Zhao, and Q. Xie, “A method to reduce ZnO grain resistance and im­prove the intergranular layer resistance by Ca2+ and Al3+ co-doping,” Materials Science in Semicon­ductor Processing, vol. 128, 2021. https://doi.org/10.1016/j.mssp.2021.105768 29. W. Deng, Z. Q. Lei, J. L. Zhao, Y. Q. Lu, and D. K. Xiong, “Eff ect of Fe2O3 Dopant on Electronic Den­sities and Electrical Properties of ZnO-Based Varis­tors,”Advanced Materials Research, vol. 415-417, pp. 1042-1045, 2011. https://doi.org/10.4028/www.scientific.net/AMR.415-417.1042 30. M. Peiteado, A. M. Cruz, Y. Reyes, J. De Frutos, D. G. Calatayud, and T. Jardiel, “Progressive degra­dation of high voltage ZnO commercial varistors upon Fe2O3 doping,” Ceramics International, vol. 40, no. 8, pp. 13395-13400, 2014. https://doi.org/10.1016/j.ceramint.2014.05.057 31. J. Shen et al., “Effects of Fe and Al co-doping on the leakage current density and clamp voltage ratio of ZnO varistor,” Journal of Alloys and Com­pounds, vol. 747, pp. 1018-1026, 2018. https://doi.org/10.1016/j.jallcom.2018.03.106 Slavko Bernik et al.; Informacije Midem, Vol. 52, No. 4(2022), 215 – 226 32. Z. Peng et al., “Influence of Fe2O3 doping on mi-crostructural and electrical properties of ZnO– Pr6O11 based varistor ceramic materials,” Journal of Alloys and Compounds, vol. 508, no. 2, pp. 494­499, 2010. https://doi.org/10.1016/j.jallcom.2010.08.100 Copyright © 2022 by the Authors. This is an open access article dis­tributed under the Creative Com­mons Attribution (CC BY) License (https://creativecom-mons.org/licenses/by/4.0/), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Arrived: 29. 09. 2022 Accepted: 09. 11. 2022 Original scientific paper https://doi.org/10.33180/InfMIDEM2022.403 Electronic Components and Materials Vol. 52, No. 4(2022), 227 – 237 VD-EXCCII Based Mixed Mode Biquadratic Universal Filter Employing Grounded Capacitors Ramesh Mishra1, Ganga Ram Mishra2, Mohammad Faseehuddin3, Jahariah Sampe4 1Department of Electronics, Dr. Rammanohar Lohia Avadh University 2Department of Physics & Electronics, Dr. Rammanohar Lohia Avadh University 3Department of Electronics and Telecommunication, Symbiosis Institute of Technology (SIT), Symbiosis International University (SIU), Lavale, Mulshi, Pune, Maharashtra, India 4Institute of Microengineering and Nanoelectronics (IMEN), Universiti Kebangsaan Malaysia (UKM), Level 4 MINES Lab, UKM 43600 Bangi, Selangor, Malaysia Abstract: A recently developed active building block (ABB), namely Voltage Differencing Extra X current conveyor (VD-EXCCII), is used to design an electronically tunable mixed-mode universal filter. The filter provides low pass (LP), high pass (HP), band pass (BP), band reject (BR), and all pass (AP) responses in voltage-mode (VM), current-mode (CM), trans-impedance-mode (TIM), and trans-admittance-mode (TAM). The filter employs two VD-EXCCIIs, three resistors, and two capacitors. The attractive features of the filters include: (i) ability to operate in all four modes, (ii) use of grounded capacitors, (iii) tunability of Q factor independent of pole frequency, (iv) low output impedance for VM and TIM mode, (v) high output impedance explicit current output for CM and TAM and (vi) no requirement for double/negative input signals (voltage/current) for response realization. The VD-EXCCII and its layout is designed and validated in Cadence virtuoso using 0.18µm process design kit (PDK) at supply voltage of ±1.25 V. The operation of filter is examined at 16.42 MHz frequency. The non-ideal gain and sensitivity analysis is also carried out to study the effect of process and components spread on the filter performance. The obtained results bear a close resemblance with the theoretical findings. Keywords: Analog signal processing; mixed-mode filter; current conveyor; VD-EXCCII VD-EXCCII na osnovi mešanega nacina bikvadraticnega univerzalnega filtra, ki uporablja ozemljene kondenzatorje Izvlecek: Nedavno razviti aktivni gradnik (ABB), in sicer napetostni diferencni pretvornik Extra X (VD-EXCCII), se uporablja za zasnovo elektronsko nastavljivega univerzalnega filtra z mešanim nacinom delovanja. Filter omogoca odzive nizke prepustnosti (LP), visoke prepustnosti (HP), pasovne prepustnosti (BP), pasovne zavrnitve (BR) in vse prepustnosti (AP) v napetostnem (VM), tokovnem (CM), trans-impedancnem (TIM) in trans-admitancnem (TAM) nacinu. Filter uporablja dva VD-EXCCII, tri upore in dva kondenzatorja. Privlacne lastnosti filtrov so: (i) možnost delovanja v vseh štirih nacinih, (ii) uporaba ozemljenih kondenzatorjev, (iii) nastavljivost faktorja Q neodvisno od frekvence polov, (iv) nizka izhodna impedanca za nacin VM in TIM, (v) visoka izhodna impedanca z eksplicitnim izhodnim tokom za CM in TAM ter (vi) ni potrebe po dvojnih/negativnih vhodnih signalih (napetost/tok) za realizacijo odziva. VD-EXCCII in njegova postavitev sta zasnovana in potrjena v Cadence virtuoso z uporabo 0,18µm kompleta za nacrtovanje procesov (PDK) pri napajalni napetosti ±1,25 V. Delovanje filtra je preverjeno pri frekvenci 16,42 MHz. Izvedena je tudi analiza neidealnega ojacanja in obcutljivosti, da bi preucili vpliv razširjenosti procesa in komponent na delovanje filtra. Dobljeni rezultati so zelo podobni teoreticnim ugotovitvam. Kljucne besede: Obdelava analognih signalov; filter v mešanem nacinu; tokovni transporter; VD-EXCCII * Corresponding Author’s e-mail: rameshmishra1985@gmail.com How to cite: R. Mishra et al., “VD-EXCCII Based Mixed Mode Biquadratic Universal Filter EmployingGrounded Capacitors", Inf. Midem-J. Microelectron. Electron. Compon. Mater., Vol. 52, No. 4(2022), pp. 227–237 R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 1 Introduction Recently, the Current-mode active building blocks (ABBs) are widely used for analog signal processing applications. The CM ABBs show greater linearity, wide bandwidth, simple structure, low power consump­tion, and enhanced dynamic range [1-6]. The universal frequency filters find a broad application spectrum in communication, control, instrumentation, data acqui­sition systems (in the analog front-end), biomedical signal processing, oscillator design, etc. [1, 2, 5, 7]. With the advancement of technology, mixed-mode systems are being developed, requiring interaction between CM and VM circuits. This task can be accomplished by TAM and TIM filters that not only perform signal pro­cessing, but also provide interfacing between VM and CM systems by acting as a bridge [8, 9]. The develop­ment of mixed-mode universal filters that can provide LP, HP, BP, BR, and AP filter functions in CM, VM, TAM, and TIM modes of operation is needed for mixed-signal system implementation. Several designs of single-input multi-output (SIMO) and multi-input single-output (MISO) mixed-mode fil­ters have been proposed in the literature that employ CM ABBs. A detailed comparison of the MISO filters with the proposed design is presented in Table 1, based on the following important measures of comparison: (i) number of CM-ABBs employed, (ii) the number of passive components needed, (iii) employment of all Table 1: Comparative study of the state-of-the-art MISO Mixed mode filter designs with the proposed filter References Mode of Operation (i) (ii) (iii) (iv) (v) (vi) (vii) (viii) (ix) (x) (xi) [10] MISO 6-OTA 2C Yes N.A. No No No Yes Yes Yes - [6] MISO 7-CCII 2C+8R No Yes No Yes No Yes Yes No - [11] MISO 3-CCII 3C+4R+ 2-switch No No No Yes No Yes Yes No - [12] MISO 4-OTA 2C Yes N.A. No No No Yes Yes Yes 2.25 MHz [17] MISO 5-OTA 2C Yes N.A. No Yes No Yes No Yes 1.59 MHz [35] MISO 2-MOCCCII 2C+2R No Yes Yes Yes No Yes Yes Yes 1.27 MHz [38] MISO CFOA 2C+3R No No Yes No No Yes No No 12.7MHz [20] MISO 4-MOCCCII 2C Yes N.A. No Yes Yes Yes No Yes - [37] MISO 1-FDCCII 2C+2R No Yes No No No Yes Yes No 10 MHz [43] MISO 2-VDTA 2C Yes N.A Yes No No Yes Yes Yes 1 MHz [30] MISO 1-FDCCII+1­DDCC 2C+6R No Yes Yes Yes No Yes No No 1.59 MHz [29] MISO 5-DVCC 2C+5R Yes Yes Yes Yes No Yes Yes No 1MHz [45] MISO 4-CCII 2C+4R Yes Yes Yes No No Yes Yes No 31.8 MHz [26] MISO 5-OTA 2C Yes Yes Yes Yes No Yes Yes Yes 3.390 MHz [27] MISO 3-DDCC 2C+4R No Yes No Yes No Yes Yes No 3.978 MHz [45] MISO 1-EXCCCII 2C No N.A. Yes No No Yes No Yes 23 MHz [47] MISO 4-ZC-CCTA 2C No N.A. No Yes No Yes No Yes 7.5 MHz [40] MISO 2-EXCCTA 2C+4R No Yes Yes Yes No Yes Yes Yes 7.622 MHz [42] MISO 2-VD-DVCC 2C+3R No Yes Yes Yes No Yes Yes Yes 5.305 MHz [44] MISO 2-VDBA 2C+2R No Yes Yes No Yes No Yes Yes 1.52 MHz [41] MISO 1-VD-EXCCII 2C+2R No Yes Yes Yes No Yes Yes Yes 8.08 MHz This Works MISO 2-VD-EXCCII 2C+3R No Yes Yes Yes No Yes Yes Yes 16.42 MHz *Full nomenclature of the mentioned ABBs in Tables 1 in alphabetical order: CCII: Second-generation current con­veyor, CFOA: Current feedback operational amplifier, DDCC: Differential difference current conveyor, DVCC: Differen­tial voltage current conveyor, EXCCCII: Extra x current controlled current conveyor, EXCCTA: Extra x Current conveyor transconductance amplifier, FDCCII: Fully differential second-generation current conveyor, MOCCCII: Multi output cur­rent controlled current conveyor, OTA: Operational transconductance amplifier, VD-DVCC: Voltage differencing differ­ential voltage current conveyor, VDBA: Voltage differencing buffered amplifier, VDTA: Voltage differencing transcon­ductance amplifier, ZC-CCTA: Z copy-current conveyor transconductance amplifier. **N.A.-Not applicable R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 grounded passive components, (iv) no requirement for 2 Voltage differencing extra X current resistive matching except for obtaining AP response, (v) provision to control quality factor (Q) independent conveyor (VD-EXCCII) of the centre frequency, (vi) ability to provide all five filter responses in all four modes of operation, (vii) low output impedance for VM and TIM modes, (viii) avail­ability of explicit current output in CM and TAM, (ix) no requirement for double/negative input signals (volt­age/current), (x) inbuilt tunability, and (xi) test frequen­cy. The MISO filter structures [3, 6, 8-29] employ ABBs in excess of two. The filter structures in [3, 6, 9, 11, 18, 21, 27, 29-32] require more than five passive compo­nents. The filter designs in [3, 6, 9, 11, 18, 21, 27, 30-42] do not employ all grounded passive components. The filters in [3, 6, 9-12, 16, 17, 20, 23, 24, 27, 30, 33, 36-39] do not provide quality factor tuning independent of frequency. The filter structures [6, 8, 10, 12-14, 16, 19, 21, 23, 28, 32, 33, 36-38, 43, 44] do not provide all five filter responses in VM, CM, TAM, and TIM operation. The filter structures [3, 6, 9, 11, 14, 18, 21, 27-34, 37-39] lack inbuilt tunability. The literature survey shows that a limited number of truly mixed-mode filters are avail­able, and additional novel mixed-mode filter structures are needed to fill this technological void. In this research, Voltage Differencing Extra X current conveyor (VD-EXCCII) is utilized to design mixed-mode filters. The design requires two VD-EXCCIIs, two capaci­tors, and three resistors. The striking features of the proposed filter are: (i) ability to work in all four modes of operation, (ii) provision for inbuilt tunability, (iii) the filter enjoy low active and passive sensitivities, and (iv) use of all grounded capacitors. Besides these, the filters enjoy all the properties (iv-x) mentioned in Table 2. The design of the VD-EXCCII is done in Cadence Virtuoso using 0.18µm PDK. The simulation results are in close agreement with the theoretical predictions. The proposed Voltage Differencing Extra X current conveyor (VD-EXCCII) is derived by connecting extra X second generation current conveyor (EXCCII) [46] and operational transconductance amplifier (OTA). The first stage comprises OTA followed by the CCII with two cur­rent input terminals. The developed active element has characteristics of CCII and tunable OTA in one structure. The voltage-current (V-I) characteristics of the devel­oped VD-EXCCII are presented in Equations (2.1-2.4) and the block diagram is presented in Fig. 1. I  I  I  g  V V , (2.1) W WC  WC  mP N VXP  VXN VW , (2.2) I  I I , (2.3) XP ZP  ZP  I  I I (2.4) XN ZN  ZN  . Figure 2: CMOS implementation of VD-EXCCII R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 The CMOS implementation of VD-EXCCII is given in Figure 2. The first stage consists of OTA MOS transistors (M1-M14). The output current of the OTA depends on the voltage difference (VP – VN). Assuming that all tran­sistors are operating in saturation region and transis­tors (M1-M2) have equal width to length ratio, the out­put current is given by Equation 2.5. The second stage is made up of hybrid voltage and current followers (M15­M44). The voltage developed at node W is transferred to nodes XP and XN. In the same way, the input current from X node is transferred to Z and Z. Furthermore, PP+P- the input current from XN node is transferred to ZN+ and ZN-. The current following in ZN and ZP terminals are in­dependent of each other. The class AB output stage is utilized in the output stage as it is suitable for low volt­age operation and better dynamic range [2]. I W  I WC  I WC  gmi  VP  VN  2 IBias Ki  VP  VN  , (2.5) Ki = µCoxW/2L, (i = 1, 2) is the transconductance param­eter, W is the effective channel width, L is the effective length of the channel, Cox is the gate oxide capacitance per unit area and µ is the carrier mobility. 3 Proposed electronically tunable mixed-mode universal filter The proposed filter, as shown in Fig. 3, requires two VD-EXCCIIs, two capacitors, and three resistors. The filter offers low output impedance for VM and TIM mode of operation. In addition, the CM and TAM responses are available from explicit high impedance terminals. Fur­thermore, the capacitors are connected to high imped­ance terminals to absorb the parasitics associated with the terminals. Among the three resistors, two are con­nected to the low resistance X terminals to accommo­date the parasitic resistance. The main drawback of the filter is the use of two floating resistors, but given the advantages of the filter, this can be accommodated. Moreover, floating resistors can be easily implemented in CMOS technology. The important features of the fil­ter include: (i) ability to provide all five filter responses in all four modes of operation, (ii) employment of a minimum number of passive components, (iii) use of grounded capacitors, (iv) no requirement for resistive matching except for AP response, (v) low output im­pedance in VM and TIM configuration, (vi) no need for capacitive matching, (vii) availability of explicit current output in CM and TAM, (viii) no requirement for double/ negative input signals (voltage/current), (xi) independ­ent control of Q and f0 and (xi) inbuilt tunability. The operation of the filter in all modes is explained below. Figure 3: Proposed Mixed-mode Filter 3.1 Operation in VM and TAM mode In this mode of operation, the inputs currents (I1 – I3) are set to zero. The filter is excited with input voltages (V1 – V3) as per the sequence given in Table 2. The transfer functions for VM/TAM and expressions for quality fac­tor and pole frequency are given in Equations (3.1-3.4). 2 sCCRRV  sCg RRV  Rg RgV 12231 1 m 1213 1 m 12 m 22 out VM 2 (3.1) V   sCCRR  sCg RR Rg Rg 1223 1 m 131 1 m 12 m 2 For all pass response, a simple resistive matching of (R3 = R2) is required, which is easy to achieve. (3.2) 1 ggR m 1 m21 f 0  , (3.3) 2 CC R 123 Cg 2 m 2 QR 2  . (3.4) Cg RR 1 m 113 Table 2: Excitation Sequence for VM and TAM 3.2 Operation in CM and TIM mode: In this mode of operation, all input voltages (V1 – V3) are set to zero. The input currents (I1 – I3) are applied R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 according to Table 3. The transfer functions for TIM and CM are given in Equations (3.5) and (3.6). 2  sCC R RI  sCRg RI  g RI  12231 13 m 123 m 122 Vout TIM   R 1  2  (3.5) sCC R R  sCg RR  Rg Rg  1223 1 m 131 1 m 12 m 2  V out TIM  I  out CM  R 2 (3.6) 1  212231  sC R g 13 m 12RI 3  gm 12RI 2  R sCCRRI   2  R sCCRR  sCg RR  Rg Rg 2  1223 1 m 131 1 m 12 m 2  In Equation 3.6 for R1 = R2 the filter gain constants are 1 HoHP  1 , HoLP  , HoBP 1 by adjusting these pa- Rg 1 m 2 rameters the filter gain can be adjusted. Table 3: Input current excitation sequence 4 Non-ideal gain and sensitivity analysis The non-ideal effects that influence the response of the VD-EXCCII are the frequency-dependent non-ideal current (a), voltage (ß), and transconductance P/N, a’P/NP/Ntransfer (., .›) gains. These non-ideal gains result in a change in the current and voltage signals during trans­fer leading to an undesired response. Taking into ac­count the non-ideal gains, the V-I characteristics of the VD-EXCCII in (3.1-3.4) will be modified as follows: IW = 0, V = ßVW, V = ßVW, I = a IXP, I = –a’ IXP, I = aIXN, XPPXNNZP+pZP–p ZN+N I = a’IXN, I = I = .g(V– V),I = –.’g(V– V), ZN–NWWC+mP NWC–mP N where ß = 1 – e = 1 – e = 1 – e = 1 PmvPm, ßNmvNm, aPmiPm, aNm – e = 1 – egmm, and .’ = 1 – e’gmm, for m = 1, 2, which iNm, .mmrefers to the number of VD-EXCCIIs. Here, e, e(|e|, vPmvNmvPm |e|«1) denote voltage tracking error, e,e (|e|, vNmiPmiNmiPm|eiNm|«1), denote current tracking errors, and egmm, , e›gmm (|egmm|, |e›gmm|«1) denote transconductance errors of the VD-EXCCII. The non-ideal analysis considering the effect of non-ideal current, voltage, and transconductance transfer gains is carried out for (VM, CM, TAM and TIM) con­figurations to see its effect on the transfer function, f0, and Q of the proposed filter. The modified expressions of filter transfer functions, f’ 0, and Q’ are presented in Equations (4.1) to (4.6). The procedure to perform the non-ideal analysis can be found in [42]. (4.1) (4.2) (4.3) (4.4) (4.5) (4.6) N 2 N 2 11 m113 R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 The sensitivities of .’ 0 and Q’ with respect to the non-ideal gains and passive components are given below in Equations (4.7) to (4.9). ''''''' ' ' '1 ooooooo o o o Sg  Sg  SR  S   S  S  S  SC  SC  SR  (4.7) m 1 m 21 N 1 N 121 123 2 Q . Q . Q . Q . Q . Q . Q . Q . Q . Q . S  S  S  S  S  S  S  S  S  S 1, (4.8)  C gCg RR N 1 NI 12 m 22 m 11 31 2 Q . Q . Q . SR  S  N '2  S 1 (4.9) 2 N 2 The sensitivities are low and have absolute values not higher than unity. 5 Simulation and validation To verify the proposed mixed-mode filter, it is designed and simulated in Cadence virtuoso design software. The newly proposed VD-EXCCII is designed in 0.18 Figure 4: Layout of the VD-EXCCII µm Silterra Malaysia technology at ±1.25V supply voltage. The widths and lengths of the transistors are given in Table 4. The bias current of the OTA is fixed at 120 µA resulting in a transconductance of 1.0321 µS. The layout of the VD-EXCCII Fig. 4 is drawn using the nhp and php high-performance MOS transistors from the Silterra library. The layout occupies a total area of 54.28*22.80µm2. The filter is designed for centre frequency of 16.4263 MHz and a quality factor of one by selecting passive Figure 5: VM MISO configuration: Frequency responses component as R = R = R =1 kO, C = C = 10pF and 12312 of the LP, BP, HP, and BR filter gm = 1.0321 µS. All five filter responses in VM, CM, TAM, and TIM modes are presented in Figs. 5-12. The simu­lated frequency for VM-AP is found to be 15.977 MHz leading to a 2.73% error. Table 4: Width and Length of the MOS transistors Transistors Width (µm) Length (µm) M1–M2, M5–M6 1.8 0.36 M3–M4, M7–M9 5.7 0.36 M10–M14 1.8 0.72 M15–M18 3.06 0.36 M19–M22 10 0.36 M23, M25, M27, M33, M42, M44 2.16 0.36 M24, M26, M28, M32, M34, M30, M38, M36, M41, M43 0.72 0.72 M21, M31, M35, M37 1.08 0.72 Figure 6: VM MISO configuration: Gain and phase re­sponses of the AP filter Figure 7: TAM MISO configuration: Frequency respons­es of the LP, BP, HP, and BR filter R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 Figure 11: TIM MISO configuration: Frequency re­sponses of the LP, BP, HP, and BR filter To examine the signal processing capability of the pro­posed universal filter, the transient analysis is carried out in VM mode for HP, LP, BR, and BP responses. A VM sinu­soidal signal of 100 mVp-p and 16.42 MHz frequency is applied at the input and the output is analyzed as pre­sented in Fig. 13. Similarly, a CM sinusoidal signal of 50 µA p-p and 16.42 MHz frequency is applied at the input, and the BP output in CM is observed as shown in Fig. 14. In the presented filter, the quality factor can be set independently of the pole frequency of the filter. The tunability of the quality factor is verified by analysing BP response in VM for different values of resistor R2as shown in Fig. 15. It can be inferred from the figure that the qual­ity factor of the filter can be tuned independently of the frequency. The frequency tuning is verified by varying the bias current of the OTAs and observing the CM-BP and VM-LP responses. It can be deduced that the fre­quency can be tuned without disturbing the Quality fac­tor of the filter as presented in Fig. 16-17. R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 Figure 18: VM MISO configuration: The Monte Carlo analysis result for BP response To study the effect of process spread on the perfor­mance of the designed filter, Monte Carlo analysis is carried out for 200 runs. The Monte Carlo analysis re­sults for VM-BP response are given in Fig. 18 and 19. The results for CM-AP configuration is given in Fig. 20. The total harmonic distortion (THD) of the filter for VM­BP response is plotted for different input signal values, as shown in Fig. 21. The THD plot for CM-BP and CM-LP is presented in Fig. 22. The THD remains within accept­able limits (=5%) for the appreciable input range. The input and output noise of the filter for VM-LP con­figuration is shown in Fig. 23. The input referred noise magnitude in the pass band of VM-LP is found in the range of 2.914E-08 V/Hz1/2. The magnitude of output re­ferred noise is in the range of 3.425E-08 V/Hz1/2. R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 6. Conclusion This paper presents a new VD-EXCCII based electroni­cally tunable mixed-mode filter. The filter employs two VD-EXCCIIs, three resistors, and two grounded capaci­tors. The presented MISO filter has inbuilt tunability and can realize all five filter responses in all four modes of operation (VM, CM, TAM, and TIM). Detailed theoreti­cal analysis and non-ideal gain analysis are done. The VD-EXCCII is designed in Cadence Virtuoso software and extensive simulations are carried out to examine and validate the proposed filter in all four modes of operation. The proposed filter has all the advantages mentioned in Table 2 (iv)-(x). The filter is designed for a frequency of 16.42 MHz at ±1.25 V power supply. The Monte Carlo analysis shows that the frequency devia­tion is within acceptable limits. Furthermore, the THD is within 5% for a considerable voltage/current input signal range. The simulation results are found consist­ent with the theoretical predictions. 7 Conflict of interest Authors declare no conflict of interest 8 Acknowledgement This work is funded by UKM Internal grant and PAME SDN BHD Industry grant under the grants GUP-2022­069 and RR-2022-001 respectively. 9 References 1. P. A. Mohan, Current-mode VLSI analog filters: de­sign and applications: Springer Science & Business Media, 2003. 2. G. Ferri and N. C. Guerrini, Low-voltage low-power CMOS current conveyors: Springer Science & Busi­ness Media, 2003. 3. R. Senani, “Novel mixed-mode universal biquad confi guration,” IEICE Electronics Express, vol. 2, pp. 548-553, 2005. 4. F. Mohammad, J. Sampe, S. Shireen, and S. H. M. Ali, “Minimum passive components based lossy and lossless inductor simulators employing a new active block,” AEU-International Journal of Elec­tronics and Communications, vol. 82, pp. 226-240, 2017. 5. R. Raut and M. Swamy, Modern analog filter analy­sis and design: a practical approach: John Wiley & Sons, 2010. 6. M. T. Abuelma’atti and A. Bentrcia, “A novel mixed-mode CCII-based fi lter,” Active and Passive Elec­tronic Components, vol. 27, pp. 197-205, 2004. 7. R. Senani, D. Bhaskar, and A. Singh, Current con­veyors: variants, applications and hardware imple­mentations vol. 560: Springer, 2015. 8. M. T. Abuelma’atti, “A novel mixed-mode current-controlled current-conveyor-based fi lter,” Active and passive electronic components, vol. 26, pp. 185-191, 2003. 9. M. T. Abuelma’atti, A. Bentrcia, and S. a. M. Al-Shahrani, “A novel mixed-mode current-convey­or-based fi lter,” International Journal of Electronics, vol. 91, pp. 191-197, 2004. 10. M. T. Abuelma’atti and A. Bentrcia, “A novel mixed-mode OTA-C fi lter,” Frequenz, vol. 57, pp. 157-159, 2003. 11. N. Pandey, S. K. Paul, A. Bhattacharyya, and J. SB, “A new mixed mode biquad using reduced num­ber of active and passive elements,” IEICE Electron­ics Express, vol. 3, pp. 115-121, 2006. 12. M. A. Ibrahim, “Design and analysis of a mixed-mode universal filter using dual-output opera­tional transconductance amplifi ers (DO-OTAs),” in 2008 International Conference on Computer and Communication Engineering, 2008, pp. 915-918. 13. Z. Li, “Mixed-mode universal filter using MCCCII. AEU—Int,” J. Electron. Commun, vol. 63, pp. 1072­1075, 2009. 14. S. Minaei and M. A. Ibrahim, “A mixed-mode KHN-biquad using DVCC and grounded passive ele­ments suitable for direct cascading,” International Journal of Circuit Theory and Applications, vol. 37, pp. 793-810, 2009. 15. H.-P. Chen, Y.-Z. Liao, and W.-T. Lee, “Tunable mixed-mode OTA-C universal fi lter,” Analog Inte­grated Circuits and Signal Processing, vol. 58, pp. 135-141, 2009. 16. S. Maheshwari, S. V. Singh, and D. S. Chauhan, “Electronically tunable low-voltage mixed-mode universal biquad fi lter,” IET circuits, devices & sys­tems, vol. 5, pp. 149-158, 2011. 17. C.-N. Lee, “Multiple-mode OTA-C universal biquad fi lters,” Circuits, Systems and Signal Processing, vol. R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 R. Mishra et al.; Informacije Midem, Vol. 52, No. 4(2022), 227 – 237 29, pp. 263-274, 2010. 32. J.-W. Horng, C.-M. Wu, and N. Herencsar, “Cur­ 18. W.-B. Liao and J.-C. Gu, “SIMO type universal rent-mode and transimpedance-mode universal mixed-mode biquadratic fi lter,” 2011. biquadratic filter using two current conveyors,” 19. M. Kumngern and S. Junnapiya, “Mixed-mode universal filter using OTAs,” in 2012 IEEE Interna­tional Conference on Cyber Technology in Automa­tion, Control, and Intelligent Systems (CYBER), 2012, 33. 2017. N. A. Shah and M. A. Malik, “Multifunction mixed-mode filter using FTFNs,” Analog Integrated Cir­cuits and Signal Processing, vol. 47, pp. 339-343, pp. 119-122. 2006. 20. 21. N. Pandey and S. K. Paul, “Mixed mode universal fi lter,” Journal of Circuits, Systems and Computers, vol. 22, p. 1250064, 2013. L. Wang, C. Wang, J. Zhang, X. Liang, and S. Jiang, 34. C.-N. Lee and C.-M. Chang, “Single FDCCII-based mixed-mode biquad filter with eight outputs,” AEU-International Journal of Electronics and Com­munications, vol. 63, pp. 736-742, 2009. 22. “A new mixed-mode filter based on MDDCCs,” in Seventh International Conference on Graphic and Image Processing (ICGIP 2015), 2015, p. 981717. H.-P. Chen and W.-S. Yang, “Electronically tunable 35. N. Pandey, S. K. Paul, A. Bhattacharyya, and S. Jain, “Realization of Generalized Mixed Mode Univer­sal Filter Using CCCIIs,” Journal of Active & Passive Electronic Devices, vol. 5, 2010. current controlled current conveyor transcon­ 36. S. Singh, S. Maheshwari, and D. Chauhan, “Elec­ ductance amplifier-based mixed-mode biquad­ratic filter with resistorless and grounded capaci­tors,” Applied Sciences, vol. 7, p. 244, 2017. tronically tunable current/voltage-mode univer­sal biquad fi lter using CCCCTA,” International J. of Recent Trends in Engineering and Technology, vol. 23. V. Chamnanphrai and W. Sa-ngiamvibool, “Elec­ 3, pp. 71-76, 2010. 24. tronically tunable SIMO mixed-mode universal fi lter using VDTAs,” Przeglad Elektrotechniczny, vol. 93, pp. 207-211, 2017. M. Parvizi, A. Taghizadeh, H. Mahmoodian, and Z. 37. F. Kaçar, A. Kuntman, and H. Kuntman, “Mixed-mode biquad filter employing single active ele­ment,” in 2013 IEEE 4th Latin American Symposium on Circuits and Systems (LASCAS), 2013, pp. 1-4. 25. D. Kozehkanani, “A Low-Power Mixed-Mode SIMO Universal G m–C Filter,” Journal of Circuits, Systems and Computers, vol. 26, p. 1750164, 2017. U. Cini and M. Aktan, “Dual-mode OTA based bi­ 38. E. Yuce, “Fully integrable mixed-mode universal biquad with specific application of the CFOA,” AEU-International Journal of Electronics and Com­munications, vol. 64, pp. 304-309, 2010. 26. quadratic filter suitable for current-mode applica­tions,” AEU-International Journal of Electronics and Communications, vol. 80, pp. 43-47, 2017. D. R. Bhaskar, A. Raj, and P. Kumar, “Mixed-mode universal biquad filter using OTAs,” Journal of Cir­cuits, Systems and Computers, vol. 29, p. 2050162, 39. 40. B. Chaturvedi, J. Mohan, and A. Kumar, “A new ver­satile universal biquad configuration for emerg­ing signal processing applications,” Journal of Cir­cuits, Systems and Computers, vol. 27, p. 1850196, 2018. M. I. A. Albrni, F. Mohammad, N. Herencsar, J. Sam­ 2020. pe, and S. H. M. Ali, “Novel electronically tunable 27. C.-N. Lee and W.-C. Yang, “General Mixed-Mode biquadratic mixed-mode universal fi lter capable Single-Output DDCC-based Universal Biquad Fil­ter,” Int. J. Eng. Res, vol. 9, pp. 744-749, 2020. of operating in MISO and SIMO confi gurations,” Inf. MIDEM, vol. 50, pp. 189-203, 2020. 28. T. Ettaghzouti, N. Hassen, and K. Besbes, “Novel 41. M. Faseehuddin, N. Herencsar, M. A. Albrni, and J. multi-input single-output mixed-mode universal Sampe, “Electronically tunable mixed-mode uni- filter employing second generation current con­veyor circuit,” Sensors, Circuits & Instrumentation Systems: Extended Papers, vol. 6, p. 53, 2017. versal filter employing a single active block and a minimum number of passive components,” Ap­plied Sciences, vol. 11, p. 55, 2020. 29. T. Tsukutani and N. Yabuki, “A DVCC-based mixed­ 42. M. Faseehuddin, N. Herencsar, M. A. Albrni, S. mode biquadratic circuit,” J. Electr. Eng, vol. 6, pp. Shireen, and J. Sampe, “Electronically tunable 30. 52-56, 2018. C.-N. Lee, “Independently tunable mixed-mode universal biquad filter with versatile input/output functions,” AEU-International Journal of Electron­ics and Communications, vol. 70, pp. 1006-1019, 2016. 43. mixed mode universal filter employing grounded capacitors utilizing highly versatile VD-DVCC,” Cir­cuit World, 2021. A. Yesil and F. Kaçar, “Electronically tunable resis­torless mixed mode biquad fi lters,”Radioengineer­ing, vol. 22, pp. 1016-1025, 2013. 31. C.-N. Lee, “Mixed-mode universal biquadratic fi l-ter with no need of matching conditions,” Jour­nal of Circuits, Systems and Computers, vol. 25, p. 44. N. Roongmuanpha, M. Faseehuddin, N. Herencsar, and W. Tangsrirat, “Tunable Mixed-Mode Voltage Diff erencing Buff ered Amplifi er-Based Universal 1650106, 2016. Filter with Independently High-Q Factor Control­lability,” Applied Sciences, vol. 11, p. 9606, 2021. 45. T. Ettaghzouti, N. Hassen, and K. Besbes, “A Novel Multi-Input Single-Output Mixed-Mode Universal Filter Employing Second Generation Current Con­veyor Circuit,” Sensors, Circuits & Instrumentation Systems: Extended Papers 2017, vol. 6, p. 53, 2018. 46. S. Maheshwari, “Realization of simple electronic functions using EXCCII,”Journal of Circuits, Systems and Computers, vol. 26, p. 1750171, 2017. 47. S. V. Singh, R. S. Tomar, and M. Goswami, “A Cur­rent Tunable Mixed Mixed Mode ZC-CCTAs Based Resistor Less Universal Filter,” Journal of Circuits, Systems and Computers, vol. 30, p. 2150225, 2021. https://doi.org/10.1142/S021812662150225X. Copyright © 2022 by the Authors. This is an open access article dis­tributed under the Creative Com­mons Attribution (CC BY) License (https://creativecom-mons.org/licenses/by/4.0/), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Arrived: 04. 07. 2022 Accepted: 16. 11. 2022 Original scientific paper https://doi.org/10.33180/InfMIDEM2022.404 Electronic Components and Materials Vol. 52, No. 4(2022), 239 – 262 Mixed-mode Universal Filter Using FD-CCCTA and its Extension as Shadow Filter Divya Singh*, Sajal K. Paul Department of Electronics Engineering, Indian Institute of Technology (Indian School of Mines), Dhanbad, Jharkhand, India Abstract: This paper presents a fully differential current conveyor cascaded transconductance amplifier (FD-CCCTA), a modified FD-second generation current conveyor (FD-CCII) version. After that, a novel mixed-mode universal filter (UF) is developed employing only one FD-CCCTA. It results in all the four modes of UFs, namely current mode (CM), voltage mode (VM), transimpedance mode (TIM), and transadmittance mode (TAM). Moreover, this filter topology is extended to two mixed-mode universal shadow filters. The first shadow filter topology realizes the VM and CM universal filters. The second mixed-mode universal shadow filter realizes all four modes. The proposed shadow filters add flexibility in the orthogonal tuning of filter parameters, .0 and Q0. Further, the gain of the shadow filter can be tuned electronically. Matching constraint is not required in any of the filters. The functional verifications have been performed using TSMC 180 nm technology in cadence virtuoso spectre. Keywords: FD-CCCTA; FD-CCII; mixed-mode; shadow-filter Univerzalni filter z mešanim nacinom uporabe FD­CCCTA in njegova razširitev kot filter v senci Izvlecek: Clanek predstavlja popolnoma diferencialni kaskadni transkondukcijski ojacevalnik (FD-CCCTA), modificirano razlicico tokovnega transporterja FD druge generacije (FD-CCII). Razvit nov univerzalni filter (UF) z mešanim nacinom delovanja, ki uporablja samo en FD-CCCTA. Rezultat so vsi štirje nacini UF, in sicer tokovni nacin (CM), napetostni nacin (VM), transimpedancni nacin (TIM) in transadmitancni nacin (TAM). Poleg tega je ta topologija filtra razširjena na dva univerzalna sencna filtra z mešanim nacinom delovanja. Prva topologija sencnega filtra izvaja univerzalna filtra VM in CM. Drugi univerzalni filter v senci z mešanim nacinom delovanja omogoca vse štiri nacine delovanja. Predlagani filtri v senci povecujejo fleksibilnost pri ortogonalnem nastavljanju parametrov filtra, ._0 in Q_0. Poleg tega je mogoce elektronsko nastaviti ojacenje filtra v senci. Pri nobenem od filtrov ni potrebna omejitev ujemanja. Funkcionalna preverjanja so bila izvedena s 180 nm tehnologijo TSMC v programu cadence virtuoso spectre. Kljucne besede: FD-CCCTA; FD-CCII; mešani nacin; filter senc * Corresponding Author’s e-mail: divs0508singh@gmail.com 1 Introduction Mixed-mode filters with all the responses of current-mode (CM) (both the input and output as a current), voltage- mode (VM) (both the input and output as a voltage), transimpedance-mode (TIM) (input as a cur­rent and output as a voltage), and transadmittance-mode (TAM) (input as a voltage and output as a cur­rent) are very much desirable in the analog signal processing, communication, and instrumentation [1]. At the same time, TAM and TIM filters play a vital role in the circuits which intends to connect the current mode circuits to the voltage mode circuits and vice-versa. TAM and TIM avoid the unnecessary circuitry requirement during V-I interfacing and the improve­ment in the effectiveness of the circuit. It concludes that the mixed-mode filters with all the four modes present in the same topology provide ample flexibility for analog circuit design. Few single-input-multiple-output (SIMO) mixed-mode universal filter topolo­gies are available in the literature. SIMO [2] has got How to cite: D.Singh et al., “Mixed-mode Universal Filter Using FD-CCCTA and its Extension as Shadow Filter", Inf. Midem-J. Microelectron. Electron. Compon. Mater., Vol. 52, No. 4(2022), pp. 239–262 D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 an advantage over single-input-single-output (SISO), multiple-input-multiple-output (MIMO), and multiple input single output (MISO) because of the availability of all the responses simultaneously. The topology [3] uses two fully differential second-generation current conveyors (FDCCIIs) with floating passive elements as four resistors and two capacitors to realize the UF in VM and TIM while multifunction filters in TAM and CM. It also requires matching components. Another three CC-CCTAs based mixed-mode topology [4] provide LP, BP, and HP responses in CM and TIM, whereas LP, BP, HP, and BR responses in VM, and UF in TAM. In [5], four OTAs are used to realize LP, BP, and HP responses in TAM, TIM, CM, and LP responses in VM. The topology [6] uses six OTAs, one resistor, and two capacitors to realize UF in VM and TIM, whereas realizing BP and HP responses in TAM and BP, HP, and BR responses in CM. It possesses floating passive components and lacks independent tuning of filter parameters. Three differential differ­ence current conveyors (DDCCs), four resistors, and two capacitors with matching components and float­ing passive elements realize UF in VM and TIM while LP, BP, and HP in TAM and CM in [7]. In [8], three dual voltage current conveyors (DVCCs), six MOSs, and two capacitors are used to realize LP, BP, and HP in CM and TAM, whereas LP, BP in TIM while LP, BP, and BR in VM with matching components requirement and no elec­tronic tunability. Mixed-mode universal filter reported in [9] uses six operational transconductance amplifiers (OTAs). The topology [10] requires three four-terminal floating nullors (FTFNs), three resistors, and two capaci­tors to realize the low pass (LP), band pass (BP), and high pass (HP) simultaneously in all the modes without independent and electronic tuning. In [11], one FDC­CII, three resistors, and two capacitors realize UF in VM and TIM, whereas BP and HP responses in TAM, and BP, HP, and, band reject (BR) responses in CM. It possesses floating passive elements and does not have an elec­tronic tunability feature. Simultaneously three respons­es (LP, BP, and HP) in all the modes are proposed in [12] with the use of five multiple-output current-controlled conveyors (MCCCIIs) and two capacitors. Reports of several shadow filters using various building blocks are available in the literature. However, among them, the majority are either in VM [13-20] or CM [21­ 28] and only one topology [29] is of TAM and TIM. To the best of the authors’ knowledge, there is no SIMO mixed-mode universal shadow filter report. The paper aims to present a novel mixed-mode uni­versal filter employing only one FD-CCCTA. Further, this filter topology is extended to a universal shadow filter for all four modes to enhance the tunability and independent variation of . and Q and provide tunable gains. The proposed circuit exhibits the following ad­vantages: least number of active building blocks, no floating components, the simultaneous realization of various responses, no need for component matching constraints, and electronic and independent tuning of filter parameters, including gains. Moreover, the pro­posed circuit is the first SIMO mixed-mode universal shadow filter to the best of the authors’ knowledge. This paper consists of six sections. The introduction is given in Section 1, followed by section 2, which de­tails the active building block FD-CCCTA. Section 3 de­scribes the proposed circuit configuration, and Section 4 describes the non-ideality analysis. Section 5 com­pares the available literature, followed by Section 6, which discusses the functional verification. 2 Fully differential current conveyor cascaded transconductance amplifier (FD-CCCTA) FD-CCCTA is a modified version of a fully differential second-generation current conveyor (FD-CCII) [30]. The symbol of FD-CCCTA is shown in Fig. 1, and its CMOS-based internal structure is shown in Fig. 2. It consists of six input terminals in the form of X as a low impedance terminal and Y as a high impedance terminal, while four output terminals in the form of Z and O as high impedance terminals. FD-CCCTA is designed using FD­CCII and transconductance amplifier (TA), where TA is used in the cascaded form, and therefore O terminal can be increased as per the requirement. D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262  IY 1  0 00000 0 0 00  VY 1      IY 2 000000 0 000 V 2    Y   IY 3  0 00000 0 0 00  VY 3      I 0 00000 0 0 00 V  Y 4   Y 4   V  1  11000 0 000  I  XA XA     (1)  VXB   110100 0 0 00  I XB      I 0 00010 0 0 00 V ZA ZA      IZB  0 00001 0 0 00  VZB      IOA 1 0 0 0000 g 0 00 VOA 1    mA 1   I  0 0 0000 0 g 00  V   OB 1  mB 1  OB 1  Where g and g are the transconductances of the mA1mB1 transconductance amplifiers (TAs) connected at the ZA and ZB, respectively, can be expressed as: W g  µC I , mA 1 n ox  A 1  L  M , M 37 38 (2) W and g  µC I mB 1 n ox  B 1  L  M , M 41 42 The aspect ratios used for transistors of Fig. 2 are given in Table 1, and the performance parameters of FD-CC­CTA are shown in Table 2. Table 1: Aspect ratios of MOS Transistors of Fig. 2. MOS Transistors W(µm)/L(µm) M1-6 4.5/0.36 M7, 8, 9, 13 36/0.36 M10, 11, 12, 24 9/0.36 M14, 15, 18, 19, 25, 29, 30, 33, 34 18/0.18 M16, 17, 20, 21, 26, 31, 32, 35, 36 4.5/0.18 M22, 23, 27, 28 0.36/0.36 M37-44 10.8/0.36 Table 2: Performance parameters of FD-CCCTA. Performance Parameters Value Supply Voltage ± 1.2 V Power Consumption 1.9 mW Parasitics at Y port (RY, CY) 2.131 MO Parasitics at X port (RX) 38.52 O Parasitics at ZA port (RZA, CZA) 1.22 MO, 28 fF Parasitics at ZB port (RZB, CZB) 4.4 MO, 12 fF Parasitics at O1 port (RO1, CO1) 2.54 MO, 3.42 fF Parasitics at O2 port (RO2, CO2) 2.3 MO, 3.24 fF Linear variation of IZ over IX 380 µA to 500 µA Linear variation of VX over VY -1.04 V to 1.04 V Bandwidth of IZ/IX 1.4 GHz Bandwidth of IO1/IX 73.1 MHz Bandwidth of IO2/IX 73.1 MHz It may be noted in the following section that the mixed-mode filter uses one FD-CCCTA; however, its extended shadow filter uses two FD-CCCTAs. Hence to distinguish the similar mathematical and non-mathematical sym­bols concerning both the blocks in the shadow filter, superscripts (1) and (2) have been used throughout the ..1 .. paper, such as and 1 for the first FD-CCCTA gmA1gmB1 22 .... and and for second FD-CCCTA. gmA1gmB1 3 Proposed circuit configuration The mixed-mode universal filter (also called non-shad­ow filter) is presented in section 3.1, followed by sec­tion 3.2, wherein two mixed-mode shadow filters are discussed. Figure 2: The CMOS-based internal structure of FD-CCCTA. D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 3.1 Mixed-mode universal filter The proposed mixed-mode universal filter, depicted in Fig. 3, consists of one FD-CCCTA, three capacitors, and one resistor. The FD-CCCTA being used in the filter is shown in Fig. 2 with the introduction of additional ...... O , BcBcBc 11 11 11 O O terminals such as , and , which are c cc 1 sC g  3mB1  (4) Ds  2 sCC 13  (5) Ds  The addition of -V and V results in V, while the ad- HPLPBR VBP Vin VHP Vin dition of -V, V, and V results in the V using another HPBPLPAP ...... OB. The OB and OB are the outputs of the other TAs. These TAs are connected in the cascaded, such as the input of the first TA is connected 11 2   sCC  gg V  13 mA 1 mB 1  ....O are 180 degrees phase-shifted 12 BR Z.. terminal to get the O .., the input of the B O.. terminal to get the BV AP O .., and similarly, the B.. terminalis obtained B 12 B 13 ...... O and O , respectively. At the same time, BBBc ...... 1 OO and -, re-2 12 11 Bc BB 13 .... 11 BcBc 13 O .., while -O.. is 180-degree phase-shifted BB 1113 13 12 12 1113 1 12 1112 copy terminals of the voltage summer (not shown).  (6) (7) to the Ds  Vin B second TA is connected to the 1 11 2    1331 11 sCC  sC g  g g mB mA mB  O Ds  Vin .OB . The and - Transimpedance Mode (TIM) [with V = 0, I = 0]: inin2 O of 11   V ggR LP mA 1 mB 1 in  (8) .O and are the copies of the Ds  I in O O 1  spectively. Similarly, and are the copies V sCg R BP 3mB1in  (9) in Ds  c I of the 2 V sCCR HP 13in O ... Thus, for obtaining the transfer functions in B all four modes, such as VM, CM, TAM, and TIM, two in­ 13 to  (10) Ds  I in put currents, I = I= I, are used, and a resistor (R) is in1in2 inin used for TAM and TIM. Figure 3: Proposed mixed-mode universal filter. The routine analysis of the circuit in Fig. 3 results in the following transfers functions: Voltage Mode (VM) [with I= I= 0, R= . (i.e. in1 in2 in Removed)]:  1  V gg 1 LP mA 1 mB 1  (3) in Ds V  The addition of -V and V results in V, while the ad- HPLPBR dition of -V, V, and V results in the V using another HPBPLPAP voltage summer (not shown) as follows: 2   sCC  g 1 g 1 R V  13 mA 1 mB 1  in BR  (11)  Iin Ds 1  2   11  sCC  sC g  gg  R VAP 13 3 mB 1 mA 1 mB 1 in  (12) Iin Ds  Where, 2   11 Ds  s CC  sC g 1 gg   (13) 13 3 mB 1 mA 1 mB 1 The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  gg Cg 1 mA 1 mB 11 mB 3 o  , Qo   CC Cg 1 13 3 mB 2 (14) 1 gmB 2  and BW C 1 D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 The sensitivity analysis of .o, Qo and BW using (14) re­sults in: oo oo S  1  S   1, SC  SC  1 1 gg 13 mA 1 mB 12 2 QQ QQ oo oo S  S  1 , S  S  1 1  1 C  C g 1 g 3 mB 32 mB 22 BW BW S  1, SC  1 1 g 1 mB 2 Current Mode (CM) [with Vin = 0, Iin1 = 0, Rin = 0]:  1  I gg 1 LP mB 2 mB 3  (15) Iin Ds  1 I sCg  BP 2mB2  (16) in Ds I  2 I sCC HP 12  (17) in Ds I  2   I sCC  g 1 g 1 BR 12 mB 2 mB 3  (18) in Ds I  2   11 I sCC  sC g 1  gg  AP 12 2 mB 2 mB 2 mB 3  (19) Iin Ds  Transadmittance Mode (TAM) [with Iin1 = 0]: Ig 1 g 1   LP mB 2 mB 3  (20)  * R V Ds in in 1 I sCg  BP 2 mB 2  (21) V Ds * R  in in 2 I sCC HP 12  (22)  * R V Ds in in 2   I sCC  g 1 g 1 BR 12 mB 2 mB 3  (23)  * R V Ds in in 1 11 2   I sCC  sC g  gg AP 12 2 mB 2 mB 2 mB 3  (24)  * R V Ds in in Where, 2   11 Ds  s CC  sC g 1 gg  (25)  12 2 mB 2 mB 2 mB 3 The pole frequency (.o), quality factor (Qo) and band­width (BW) are: 11 1   gg Cg mB 2 mB 31 mB 3 o  CC , Qo  Cg  1 12 2 mB 2 (26) 1 gmB 2  and BW C 1 The sensitivity analysis of .o, Qo and BW using (26) re­sults in: oo oo S  1  S   1, SC  S  1, 1 C gg 21 22 mB 2 mB 3 QQ QQ oo oo S  1  SC  1 , S   S  1 1 C g 1 g 2 mB 32 mB 22 BW BW S  1, SC  1 1 g 1 mB 2 The above equation indicates that the .o, Qo, and BW are electronically tunable by bias currents because of .... 1 and 1 . Sensitivity of the parameters of eqn. gmB2gmB3 (26) are found within the unity. 3.2 Mixed-mode shadow Filter Shadow filter also known as frequency agile filter, a recently introduced filters is shown in Fig. 4 [31]. The inclusion of an additional external amplifier in the feed­back of the basic filter gives the structure of shadow filter. The introduction of gain (A) of this external ampli­fier in the filter parameters improves the tunability and eases in frequency agility in comparison of the conven­tional tuning technique. Figure 4: Scheme of the shadow filter [31]. In line with Fig. 4, block diagram for the implementa­tion of mixed-mode shadow filter is shown in Fig. 5. Combination of voltage and current signals at the in­put as well as at the output form all the modes such as VM, CM, TIM, and TAM. Two amplifiers with gains A1 and A multiplied with the V and I are fed-back to 2APBP the voltage and current input signals, respectively. In this section, two topologies are proposed for mixed-mode shadow filters using basic mixed mode UF of Fig. 3. The first topology realizes VM and CM universal fil­ters, and the second topology realizes all four modes of D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Figure 5: Block diagram for the implementation of mixed-mode shadow filter. mixed-mode filters. Both the topologies have been im­plemented on the basis of the structure shown in Fig. 5. 3.2.1 First shadow filter (VM and CM) The proposed shadow filter in the VM and CM, as shown in Fig. 6, consists of the above mixed-mode universal filter of Fig. 3 along with a second FD-CCCTA block and two variable resistors (R1 & R2) consisting of MOSs. VBP ..2 .. is given to Y2 and Y42 terminals for the sake of the 2 VM shadow filter, and IBP is given to the XA.. terminal for the CM shadow filter. The second FD-CCCTA in Fig. 6 aims to create two amplifiers, A1 and A2, in the feedback loop [32] of the previous mixed-mode filter (Fig. 3) to obtain the VM and CM shadow filters. By routine analy­sis of Fig. 6, it is shown in eqn. (38) that 2  2 A gR and A  gR 1 mB 1 2 mA 2 The value of MOS resistors can be adjusted with their respective bias voltages, VC1 and VC2 [33]. The equation for the resistance is: L R  (27) 2 µC W V V  ox CiT Where L and W are the channel length and channel width, . is the effective mobility, C0X is the gate oxide capacitance, and VT is the threshold voltage of the MOS transistor. The routine analysis of the circuit Fig. 6 results in the following transfer function: The port relationships of FD-CCCTA suggests: V  V  V  V XA Y 1 Y 2 Y 3 (28) andV XB  VY 1  VY 2  VY 4 Figure 6: Proposed first shadow filter. Therefore for FD-CCCTA-1, we get: 1 V V (29) XA BP 1 V  V  V V (30) HP BPY 2 LP While for FD-CCCTA-2 we get: 2  2  V  V  V V (31) XA APBPY 3 V  V 2 V (32) in APBP Again, the port relationships of FD-CCCTA-1 results:  1 I 1 OA   1 g V 1 mA ZA  1 and I 1 OB   1 g V 1 mB ZB (33) Which corresponds to: V sC LP 3   1 g V 1 mA BP and  V sC BP 1   1 g V 1 mB HP (34) Another port relationships of FD-CCCTA-1 results:  1 I OB 2   1  1 g V and I mB 2 1 OB OB 3   1 g V mB 3 OB 2 (35) Which corresponds: I BP  1  g mB 2 I HP sC 1 and I LP   1 g mB 3 I BP sC 2 (36) Whereas, port relationships of FD-CCCTA-2 results: 22 22    I  g V andI gV (37)  OA mAZA OB mBZB Hence, (35) results: D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 2  2  2 V   I  g RI and Y 3 g 2 V  OA mA 2 BP mBAP (38) R 1 It can be written as: 2  2  I  A I andV AV (39)  OA 2 BP Y 31 AP 2 2  Where A gR and A gR (40) 1 mB 12 mA 2 Eqn. (31) can be rewritten after substituting the value of 2 .. VY 3 from (39) as  V 2  V  1  A V (41) XA AP 1 BP Now, (28) can be rewritten as: 2 V  V  V V (42) HP BPXA LP Therefore, V  V  V  1  A  V V (43) HP BPAP 1 BP LP Substituting the value of VAP from eqn. (32) into eqn. (43) gives: V  1  A  V  2 VA V (44) in 1 HP BP 1 LP Whereas for the current mode (CM), the expression of currents at node N is given as: I  I  I  1  A I (45) in HP BP 2 LP Now the voltage mode transfer functions are obtained by using eqn. (34) and eqn. (44) while, current mode transfer functions are obtained by using eqn. (36) and eqn. (45) as follows: Voltage mode [with Iin = 0]: 1  1  V gg 1  A  LP mA 1 mB 11  (46) in Ds V  1  V sCg 1  A  BP 3 mB 11  (47) Vin Ds  2 V sCC  1  A  HP 13 1  (48) in Ds V  1  2   11  sCC  2 sC g  gg  13 3 mB 1 mA 1 mB 1 VAP  (49) Vin Ds  By the addition of -V and V results into V using volt- HPLPBR age summer. 11 2    sCC  gg  1  A  VBR 13 mA 1 mB 11  (50) in Ds V  Where, 2 A  g R 1 mB 1 (51) 2   11 Ds  s CC  2 AsC g 1  gg   13 13 mB 1 mA 1 mB 1 The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  gg 1Cg 1 mA 1 mB 11 mA 1 o  , Qo   CC 2 A 1 13 1 Cg 3 mB 1 (52) 1 gmB 1  and BW  2A 1 C 1 Considering C .C , eqn. (52) gives: 13  1  1  gg 1g 1 mA 1 mB 1 mA 1  , Q  oo  C 2 Ag 1 1 mB 1 (53) 1 gmB 1  and BW  2A 1C The gain of the filter can be expressed as:  1  A 1  A  A  A  1 A  ,A  ,  LPHP BR 1 BP 2 A 1 (54) AAP1 A 1 The sensitivity analysis of .o, Qo and BW using (52) re­sults in: oo oo S  1  S   1, S  S  1, 1 CC gg 13 mA 1 mB 12 2 QQ QQ oo oo S  1  S  1 , S   S  1, C 1 C g 1 g 3 mA 12 mB 12 oo SQ SRQ  2 g 1 mB BW BW BWBW S  1  1,SC  1, S   S  1 2 R g 1 g 1 mB 1 mB The above eqn. (53) indicates that the .o, Qo, and BW are electronically tunable by bias currents because of D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 ..1 .. , and 1 , and .is independently tunable by C, gmA1 gmB1o 2 while Q is independently tunable by gain A1, i.e., .. ogmB as well as R. Also, the tuning of gain is obtained by A1. 1 Current mode [with Vin = 0]: I LP I in   1  1 g g mB 2 mB 3  D s (55) I BP I in   1 sC g 2mB2  D s (56) I HP I in  2 s C C 12  D s (57) I BR I in  2  1  1 sC C  g g 12 mB 2 mB 3  D s (58) I AP I in  2  sC C 12  1  sC g 2 mB 2  D s   1  1 g g mB 2 mB 3  (59) Where,  Ds  2 s C C 12  1    1 A sC g 2  2 mB 2   1  1 g g mB 2 mB 3 and A 2  2 g R mA 2 (60) The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  g Cg mB 2 gmB 3 11 mB 13 o  , Qo   1 12 1  2 mB 2 CC  A 2  Cg (61) 1 gmB 2  and BW  1 A 2   C 1 Considering, C .C , eqn. (61) gives: 12 11   1 gg 1 g  mB 2 mB 3 mB 3  , Q  1 oo  C  1  A  g 2 mB 2 (62) 1 gmB 2  and BW  1 A 2  C The sensitivity analysis of .o, Qo and BW using (61) re­sults in:  1  C oo oo S  S  1, S  S  1, 1 C g mB 2 gmB 321 22 QQ QQ oo oo S  1  SC  1 , S   S  1, 1 C g 1 g 2 mB 3 mB 2 22 QQ oo 2 S SR  A 2  mA g 2 1A2 BW BW BWBW 2 S   1,SC  1, S   SR  A 12  2 g 1 g 2 mB 2 mA 1A The above equation indicates that the .o, Qo, and BW are electronically tunable by bias currents because of ..1 .. , and 1 , and .is independently tunable by gmB2 gmB3 o C, while Qo is independently tunable by gain A2 , i.e., .. 2 as well as R. Sensitivity analysis of all the param­ gmA2 eters resulted within the unity. 3.2.2 Second shadow filter (all four modes) The second shadow filter shown in Fig. 7 realizes all the modes, such as VM, CM, TAM, and TIM. This circuit is a slight alteration of Fig. 6 with the addition of one input resistor (R) and one more input current such that I inin1 = I = I in2in. Figure 7: Proposed second mixed mode shadow filter. The routine analysis of the circuit Fig. 7 in line with Fig. 6 results in the following transfer functions for VM, CM, TIM, and TAM: Voltage mode (VM) [with Iin1 = Iin2 = 0, Rin= .(Re­moved)]: 1  1  V gg 1  A  LP mA 1 mB 11  (63) Vin Ds  D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 1  V sCg 1  A  BP 3 mB 11  (64) in Ds V  2 V sCC  1  A  HP 13 1  (65) Vin Ds  1  2   11  sCC  sC g  gg  V 13 3 mB 1 mA 1 mB 1 AP  (66) V  in Ds By the addition of V and -V results into V using volt- HPLPBR age summer. 11 2    sCC  gg  1  A  VBR 13 mA 1 mB 11  (67) V  in Ds Transimpedance mode (TIM) [with Vin = 0, Iin2 = 0]: 1  1  V ggR  1  A  LP mA 1 mB 1 in 1  (68) in Ds I  1  V sCg R  1  A  BP 3 mB 1 in 1  (69) in Ds I  2 V sCCR  1  A  HP 13 in 1  (70) in Ds I  1 11 2   sCC  sC g  gg R V  13 3 mB 1 mA 1 mB 1  in AP  (71) in Ds I  By the addition of V and -V results into V using volt- HPLPBR age summer. 1 2  1   sCC  gg  R  1  A  VBR 13 mA 1 mB 1 in 1  (72) in Ds I  Where,  2   1  1  1 Ds  s CC  21  A sCg  gg 13 13 mB 1 mA 1 mB 1 2 and A g R (73) 1 mB 1 The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  gg 1Cg 1 mA 1 mB 11 mA 1  , Q  o CC o 21  A 1 13 13 mB 1  Cg  (74) 1 gmB 1  and BW  21 1  A  C 1 Considering, C .C , eqn. (74) gives: 13  o  1  1 gg 1 mA 1 mB C , Q o  1 21  A 1   1 g 1 mA  1 g 1 mB (75) and BW   21  A 1 1 1 gmB C The gain of the filter can be expressed as: A  LP and A  A   A A HP BR  1 ,1  BP 1 A  AP  21 A  1   1  A 1  ,  21  A 1  (76) The sensitivity analysis of .o, Qo and BW using (74) re­sults in: oo oo S  1  S   1, S  S  1, 1 CC gg 13 mA 1 mB 12 2 QQ QQ oo oo S  1  SC  1 , S   S  1, 1 C g 1 g 3 mA 12 mB 12 Qo Qo 1 S SR  A g mB 1 2 1A  1 BW BW BWBW 1 S  1,S  1, S 2  SR  A1  1 C  g 1 g 1 mB 1 mB 1A The (73) indicates that the .o, Qo, and BW are elec- .. tronically tunable by bias currents because of 1 gmA1 .. and 1 . Moreover, .is independently tunable by gmB1o C, while Qo is independently tunable by gain A1, i.e., .. 2 and R. Also, the VM and TIM shadow filter’s gain gmB1 is tunable by A1as indicated in (74). Current mode (CM) [with Vin = 0, Iin1 = 0, Rin = 0]:  1  I gg 1 LP mB 2 mB 3  (77) in Ds I  1 IBP sCg 2 mB 2  (78) Iin Ds  2 IHP sCC 12  (79) Iin Ds  2   I sCC  g 1 g 1 BR 12 mB 2 mB 3  (80) Iin Ds  D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 2   11 sCC  sC g 1  gg  I  12 2 mB 2 mB 2 mB 3  AP  (81) I  in Ds Transadmittance mode (TAM) [with Iin1 = 0]: Ig 1 g 1 LP mB 2 mB 3    (82) V Ds * R  in in 1 I sCg  BP 2 mB 2  (83)  * R V Ds in in 2 I sCC HP 12  (84) V Ds * R  in in 2   I sCC  g 1 g 1 BR 12 mB 2 mB 3  (85)  * R V Ds in in 1 11 2   sCC  sC g  gg I AP  12 2 mB 2 mB 2 mB 3   (86)  * R V Ds in in Where, 1 11 2   Ds sCC     1 A sCg  gg 12 22 mB 2 mB 2 mB 3 2 and A g R (87) 2 mA 2 The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  gg 1Cg 1 mB 2 mB 31 mB 3 , Qo   1 12 22 mB 2 o  CC  1  A  Cg (88) 1 gmB 2  and BW  1 A 2   C 1 Considering, C .C , eqn. (88) gives: 12  1  1  gg 1 mB 2 mB 31gmB 3 , Q  1 oo   C  1  A  g 2 mB 2 (89) 1 gmB 2  and BW  1 A 2  C The sensitivity analysis of .o, Qo and BW using (88) re­sults in: oo oo S  S  1, S  S  1, 11   CC gg 12 mB 2 mB 32 2 QQ QQ oo oo S  1  SC  1 , S   S  1, 1 C g 1 g 2 mB 32 mB 22 QQ 2 oo S SR  A 2  g 2 mA 1A2 BW BW BWBW A2 S   1,SC  1, S   SR  12  gg mB 21 mA 2 1A2 The equation (89) indicates that the .o, Qo, and BW are electronically tunable by bias currents because of ..1 .. and 1 . Moreover, .is independently tun­ gmB2gmB3 o able by C, while Qo is independently tunable by gain 2 .. A2 , i.e., gmA and R2. Sensitivity analysis of all the pa­rameters results within the unity magnitude. 4 Non-ideality analysis Non-ideal transfer gains and active building block parasitics will have an impact practically. Sections 4.1 discusses the effect due to non-ideal transfer gains. and section 4.2 discusses the effect due to parasitics of FD-CCCTA. 4.1 Non-ideal transfer gain of FD-CCCTA The port relationship is modified as follows when tak­ing into account the non-idealities of the voltage, cur­rent, and transconductance gains of FD-CCCTA:  I Y 1    I  Y 2     0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 00   00   Y V 1     Y V 2   I  Y 3   I  Y 4     0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 00   00    Y V 3    Y V 4     XA V 1 a     XB V   1 b  a 2 b 2 a 3 0 0 0 0 b 3 0 0 0 0 0 0 00   00   I  XA    I XB  (90)   I ZA    I ZB    I  1 OA       0 0 0 0 0 0 0 0 0 0  a 0 0 0  b 0 0 0 0 0 1 a g 1 mA 0 0 0  00  00   00    ZA V    ZB V    V  1 OA    I 1 OB     0 0 0 0 0 0 0 1 b g 1 mB 00   V   1 OB   Where . (i=1,2,3) is the voltage transfer gain be- ai tween Yand X terminals, . (i=1,2,3) is the volt­ (i) Abi age transfer gain between Yand X terminals, is (i) B the current transfer gain between IZA and IXA terminals, . is the current transfer gain between I and I ter- b ZBXB minals, . is the transconductance gain between I a1 OA1 and V, and . is the transconductance gain be­ OA1b1 tween I and V. These gain factors are found unity OB1OB1 ideally but they deviate slightly from unity practically. The transfer functions of Fig. 3 after considering the non-idealities are obtained as follows: D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Voltage Mode (VM): V LP V in   1  1 gg 1 mA 1 1 mB a b 2 1 1 a b  D s (91) V BP V in   1 sC g 3 1 mB b 2 1 B  D s (92) V HP V in  2 sC C 13b2  D s (93) V BR V in 2 sC C  13  b 2   1  1 g g 1 mA 1 1 2 1 1 mB a b a b   D s (94) V AP V in 2  1 sC C  sC g 13 b 2 3 1 mB b 2 1 B     1  1    gg 1 mA 1 1 mB a b 2 1 1 a b   D s (95) Transimpedance Mode (TIM): V LP I in   1  1 gg R 1 mA 1 1 mB a b 2 1 1 a b in  D s (96) V BP I in   1 sC g R 3 1 mB b 2 1 b in  D s (97) V HP I in  2 sC C R 13b2in  D s (98) V BR I in 2 sC C 13 b 2   R in   1  1   gg 1 mA 1 1 mB a b 2 1 1 a b   D s (99) V AP I in 2  1 sC C  sC g 13 b 2 3 1 mB b 2 1 b   R in   1  1    gg 1 mA 1 1 mB a b 2 1 1 a b   D s (100) Where,  Ds  2 s C C 13  1 sC g 3mB1 1 1a b  1  1 gg1 mA 1 1 mB a b 2 1 1 a b (101) The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  gg Cg 1 1 mB 1 1 211 b ab 1 121 mA a mA ba  , Q  oo  CC 1 13 3 mB 1 11 Cg ab (102) g 1 a mB 1 11 b  and BW C 1 Current Mode (CM):  1  I gg 1 2 mB 323 b LPmB b  (103) in Ds I  1 IBP sCg 2mB 22b  (104) in Ds I  2 I sCC HP 12  (105) in Ds I  2   I sCC  g 1 g 1 BR 12 mB 2 mBb b 323  (106) Iin Ds  2  sCC  sC g 1 b 12 2 mB 22  1  I  gg 1 (107) 2 mB 323 b AP mBb  Iin Ds  Transadmittance Mode (TAM) [with Iin1 = 0]:  1  I gg 1 2 mB 323 b LPmB b  (108) V DsR  in in 1 IBP sCg 2mB 22b  (109) DsR V  in in 2 I sCC HP 12  (110) DsR V  in in 2   I sCC  g 1 g 1 BR 12 mB 2 mBb b 323  (111) DsR V  in in 2  sCC  sC g 1 b 12 2 mB 22 11   I  gg (112) 2 mB 323 b AP mBb  V DsR  in in Where, 2   11 Ds  s CC  sC g 1 gg  b (113)  12 2 mB 22 b mB 2 mB 323 b The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  1  gg Cg 1 2 mB 323 b 1 mB 33 mBb b o  , Qo   CC Cg 1 b 12 3 mB 22 (114) g 1 mB 22  and BW b C 1 The transfer functions of Fig. 6 after considering non-idealities result: D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Voltage mode:  1  gg 1  A V mA 11  a 1 131 b  ab ab mB a 1 211 LP b 1  (115) in Ds V  1  V sCg  A  BP 31 bB a 11 a 3 11 mB 21 bb  (116) Vin Ds  2 sCC  Ab  VHP 13 b 2 a 11 a 31b 1  (117) Vin Ds  1 2  sCC  sCg B 13 b 23 mB 1 b 21  11    V  gg (118) ab 2 11 AP mA 1 mB 11 ab   in Ds V  2 sCC 13 2 ba 1     1   A gg 1 131 b (119)  a VBR mA 1 ab   mB 1 1 211 ab  b 1 V  in Ds 2 12  s CC  sC g ( Ds  131 3 mB ab b 1 b 1 11  2  1 b 2 b 3 a 13 a 2 b 31 a bb (120)  A  A ) 3 b 2 11 a 3 b 3 11 ab b  1  gg1 1 mB 1 1 211 b ab mA a The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  gg 1 ab 2 11 mA 1 mB 11 ab  , o CCb 13 1 1 1 mA 1 ba 1 Cg  21 Qo  1 (  21  a 1 b 2 b 3 Cg  1 111 b 3 mB abb  2  b a 13 ba 2 b 31 (121) 3 b 2 11  a bA  a bA ) 3 b 3 11 1 g (  2  mB 111 1 ab b 3 ab b 12  2  b a 13 ba 2 b 31  A  A ) b 3 311 a 3 b 2 11 a bb and BW  C 11 b Current mode:  1  I gg 1 2 mB 323 b LPmB b  (122) Iin Ds  1 IBP sCg 2mB 22b  (123) Iin Ds  2 I sCC HP 12  (124) in Ds I  2   I sCC  g 1 g 1 BR 12 mB 2 mBb b 323  (125) in Ds I  2   11 I  sCC  sC g 1  gg  b  12 2 mB 22 b mB 2 mB 323 b AP (126)  in Ds I  Where, 2  A  sC g  Ds  s CC 11   12 21 a 2 mB 22 ab (127)  1  gmB g 1 b 2 mB 323 b The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  gg 1 2 mB 323 b o  mB b , CC 12 1 1 Cg 1 mB  33 b (128) Qo   1  1  A  Cg 21 a 2 b a mB 22 g 1 mB 22  and BW  1 Aa  b  21 a C 1 The transfer function of Fig. 7 after considering the non-idealities are obtained as: Voltage mode (VM):  1  1 a 1 mA 1 mB 1 a b ab gg  1 211 VLP  A 131a bb 1  (129) Vin Ds  1  sCg b  Ab  VBP 3 mB 121 Ba 11 a 31b 1  (130) Vin Ds  2 sCC  Ab V 13 b 2  a 11 a 31 HP b 1  (131) in Ds V  1 2  sCC  sCg B 13 b 23 mB 1 b 21  1  1   gg  (132) V  ab 2 11 AP mA 1 mB 11 ab   Vin Ds  2 sCC 13 2 ba 1  1     1    gg A 131 b  (133) V  a  mAa b 1 BR 1 mB 1 1 211 b ab   Vin Ds  D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Transimpedance mode (TIM):  1  a 1 gg 1 R  mA 1 mB 1 in abab 1 211 V  A LP 131a bb 1  (134)  Iin Ds  1  sCg R  A  VBP 3 mB 1 in b 2 B 1 a 11 a 3 b 1b 1  (135) in Ds I  2 sCC R  A  VHP 13 in b 2 a 11 a 31bb 1  (136) in Ds I  2  sCC  sC g 1 13 b 23 mB 1 b 2 B 1  Rin 1    1  V gg (137) AP 1 mB 1 a 1 b 2 11   mA ab I  in Ds 2 sCC 13 b 2 a 1 11   in    V  gg  A 1 31 ab R (138) BR mA 1 ab   mB 1 1 211 ab  b 1  Iin Ds  Where,  2 Ds s CC 13 1  1 a 2  sC g 3 mB 1 11  (139) ab  A  A 3 211 ba 3 311 b   ab b  1  gg 1 1 mB 1 1 211 b ab mA a The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  gg 1 1 mB 1 1 211 b ab o  mA a , CC 13 1 1 mA 1 21 ba 1 Cg  Qo   1 1  a 2 Cg (140) 3 mB 1 1 11 b ab  A  A 3 211 ba 3 311 b ab b  1  1 BW  a 2 gmB 1 a 11  b   A  AC  a 3 211 bb a 3 311 bb 1 Current mode (CM):  1  I gg 1 2 mB 323 b LPmB b  (141) Iin Ds  1 IBP sCg 2mB 22b  (142) in Ds I  2 I sCC HP 12  (143) Iin Ds  2   I sCC  g 1 g 1 BR 12 mB 2 mBb b 323  (144) in Ds I  1  2   11  sCC  sC g  gg  22 mB 323 I AP 12 2 mBb mB 2 bb  (145)  Iin Ds Transadmittance mode (TAM) [with Iin1 = 0]:  1  I gg 1 2 mB 323 b LPmB b  (146)  * R V Ds in in 1 IBP sCg 2 mB  22 b  (147)  * R V Ds in in 2 I sCC HP 12  (148)  * R V Ds in in 2   I sCC  g 1 g 1 BR 12 mB 2 mBb b 323  (149) V Ds * R  in in 2   11 sCC  sC g 1  gg  I  12 2 mBb mB 2 mB 323 b  22 b AP  (150)  * R V Ds in in Where, Ds  s CC 1 ab  A  sC g  2  1 12 21 a 2 mB 22 11 (151) 2 mB 323 b   gmBg b The pole frequency (.o), quality factor (Qo) and band­width (BW) are:  1  o  gmB g 1 b 2 mB 323 b , CC 12 1 1 Cg 1 mB  33 b (152) Qo   1  1  A  Cg 21 a 2 b a mB 22 g 1 mB 22  and BW  1 Aa  b  21 a C 1 The effects caused due to non-idealities can be easily observed from the above eqns. (102, 114, 121, 128, 140, 152). However, if transfer gains are close to unity, which is normally the case, then these equations may revert into the ideal forms. 4.2 Effects of parasitics The non-ideal equivalent circuit of FD-CCCTA is shown in Fig. 8. Series resistance at XA and XB terminals are of low value. ( C || R ) , ( C || R ) , ( C || R ) , Y 1 Y 1 Y 2 Y 2 Y 3 Y 3 D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 ( C || R ) , are at Y, Y, Y, and Y terminals, respec- Y 4 Y 4 1234 t i v e l y while ( C || R ) , ( C || R ) , ZAZA ZB ZB ( C || R ) , ( C || R ) , are at Z, Z, O, and OA 1 OA 1 OB 1 OB 1 ABA1 O terminals. The values of R , R , R , R , R B1Y1 Y 2 Y 3 Y 4 ZA, RZB , ROA1 , ROB1 are high whereas CY1 , CY 2 , CY 3 , C , C , C , C , C are low. Y 4 ZAZB OA1 OB1 Figure 8: Non-ideal equivalent circuit of FD-CCCTA. The non-ideal circuit of proposed mixed-mode filter is shown in Fig. 9 where impedances are:  1 11  ,  , Z 1  C 1 R  , Z  ( C || R ) RY  1 OB 12 2 OB 2  1 1 ,  1  1  Z 3  C 3 R  , Z  ( C || R ), RY  4 OA 14 ZB ZB  1  Z  ( C || R 1) 5 ZA ZA  1 1 1  Z  C 1 in Y 2 CY  3 RY  2 RY  3 Rin  Where, 11 1 ,   ,  C  C  C  C , C  C  C , 11 OB 1 Y 12 2 OB2 ,   C  C  C 1  C 1 33 OA 1 Y 4 The routine analysis of Fig. 9 results in: Voltage Mode (VM):  1  V gg 1 LP mA 1 mB 1  (153) in Ds V  1 ,1 1   sC 3   gmB 1 V RR (154) BP  Y 4 OA 1  Vin Ds  11 11 sC  sC   1,  3,  VHP  RY 1 RO B1  RY 4 ROA 1  (155)  Vin Ds  ,1 1 ,1 1 sC  sC   1  3  RR RR  Y 1 O B1  Y 4 OA 1  (156)   1   V  gg 1 BR  mA 1 mB 1   in Ds V  ,1 1  sC 1   RR  Y 1 O B1   ,1 1  sC    3   RR   Y 4 OA 1   (157)  ,1 1  1  sC 3   gmB 1  RR  Y 4 OA 1   1    1  V  gg AP  mA 1 mB 1   in Ds V  Transimpedance Mode (TIM):  1  V gg 1 R LP mA 1 mB 1 in  (158) in Ds I  ,1   sC  1  g 1 R 3 mB 1 in V  RY 4 ROA 1  (159) BP I  in Ds ,1 1 ,1 1 sC  sC  R  1  3  in VHP  RY 1 RO B1  RY 4 ROA 1  (160) Iin Ds  ,1 1 ,1 1 sC  sC   1  3  RR RRR  Y 1 O B1  Y 4 OA 1  in (161)   1    gg 1 VBR  mA 1 mB 1  I  in Ds ,1 1  sC   1 RR  Y 1 O B1   ,1 1  sC    3   RR  R  Y 4 OA 1  in  (162)  11 1  ,   sC 3   gmB 1  RR  Y 4 OA 1     1    gg 1 VAP  mA 1 mB 1   Iin Ds  D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Where, ,1 1 ,1 1  sC 1   3  Ds  sC  RR RR  Y 1 O B1  Y 4 OA 1  11 1 (163)  sC ,   g  3 mB 1 RR  Y 4 OA 1   1  1 gg1 mB 1 mA Current Mode (CM):  1  I gg 1 LP mB 2 mB 3  (164) Iin Ds  1 1 ,   sC 2  gmB 2 IR (165) BP  O B2   in Ds I  ,1 1 ,1 sC  sC   1  2  I RRR (166) HP  Y 1 O B1  O B2   in Ds I  11 1 sC  sC   1,  2,  RR R  Y 1 O B1  O B2  (167)   I  g 1 g 1 BR mB 2 mB 3  in Ds I  11 1 sC  sC   1,  2,  RR R  Y 1 O B1  O B2  1 (168) ,   11  sC  g 1  gg  2 mB 2 mB 2 mB 3 I AP  RO B2   Iin Ds  Transadmittance Mode (TAM):  1  I gg 1 LP mB 2 mB 3  (169) V Ds * R  in in 1 1 ,   sC 2  gmB 2 IR (170) BP  O B2    * R V Ds in in ,1 1 ,1 sC  sC   1  2  IHP  RY 1 RO B1  RO B2  (171)  V Ds * R  in in ,1 1 ,1  sC 1   sC 2  RR R  Y 1 O B1  O B2  (172)  1  I  gg 1 BR mB 2 mB 3   * R V Ds in in 11 1 sC  sC   1,  2,  RR R  Y 1 O B1  O B2  1 (173) ,   11  sC  g 1  gg   2  mB 2 mB 2 mB 3 I AP  RO B2  in  * R V Ds in Where, ,1 1 ,1 Ds  sC  sC    1  2   RY 1 RO B1  RO B2  (174) 1  ,1   11  sC 2  gmB 2  gmB 2 gmB 3  RO B2  Figure 9: Non-ideal equivalent circuit of Fig. 3 with parasitics. The non-ideal circuit of proposed mixed-mode first shadow filter (Fig. 6) is shown in Fig. 10, where imped­ances are: 1 1  1  Z 6  CY  2 CY 3 || RY 12 RY  3   , 2   2   Z  C 2 C || R 2 7 Y 1 ZB Y 1 RZB   , 2   2   Z  C 2 C || R 2 ROB || R 1  8 Y 3 OB Y 3 , 2 2 Z  ( C  | R  | R 2) 9 ZA ZA , The rest of the impedances are similar to the Fig. 8 The routine analysis of Fig. 10 results VM, and CM expres­sions due to the effect of parasitics as: D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Voltage mode: 1  1  V gg 1  A  LP mA 1 mB 11  (175) in Ds V  11 ,1   sC 3   gmB 11  A 1  VBP  RY 4 ROA 1  (176)  Vin Ds  11 sC   1,  RR  Y 1 OB 1  ,1 1 (178)  sC 3   1  A 1  V RR HP  Y 4 OA 1   in Ds V  ,1 1  sC 1   RR  Y 1 OB 1   ,1 1  sC    3   RR   Y 4 OA 1   (179) ,1 1   1  sC 3   gmB 1  RR  Y 4 OA 1     1   V  gg 1 AP  mA 1 mB 1   in Ds V  ,1 1  sC   1 RR  Y 1 OB 1   ,1 1  sC   1  A   3  1  RY 4 ROA 1  (180)  11    gg  mA 1 mB 1  VBR   in Ds V  Where, 2 A  g Z 1 mB 8 ,1 1 ,1 1  sC 1   3  Ds  sC  RR RR  Y 1 O B1  Y 4 OA 1  (181) 11 1  2 AsC ,  g  1  3  mB 1 RR  Y 4 OA 1   1  1   gg I 2 Z mA 1 mB 1 XA 6 Current mode:  1  I gg 1 LP mB 2 mB 3  (182) in Ds I  1 1  sC ,  g  2 mB 2 IR (183) BP  OB 2   Iin Ds  11 1 sC  sC   1,  2,  I RRR (184) HP  Y 1 OB 1  OB 2   Iin Ds  11 1 sC  sC   1,  2,  RR R  Y 1 OB 1  OB 2  (185)   I  g 1 g 1 BR mB 2 mB 3  in Ds I  11 1 sC  sC   1,  2,  RR R  Y 1 OB 1  OB 2  (186) 1  ,1   11  sC 2  gmB 2  gmB 2 gmB 3 I AP  ROB 2   in Ds I  Where, 11 1 ,, Ds  sC  sC    1  2  RR R  Y 1 O B1  O B2  1 ,1   1 A 2  sC 2  gmB 2 (187) R  O B2   1   gg 1 mB 2 mB 3 2 and A g Z 2 mA 9 Similarly, the non-ideal circuit of proposed mixed-mode second shadow filter (Fig. 7) is shown in Fig. 11, where impedances are: 22    2  2 Z  C || R || R  , Z  ( C || R ) 7 Y 1 Y 1 in 10 ZB ZB While rest of the impedances are similar to the Fig. 8 and Fig. 9. The routine analysis of Fig. 11 results VM, TIM, CM, and TAM expressions due to the effect of para­sitics as follows: D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 ,1 1  sC 1   RR  Y 1 O B1   ,1 1  sC   3   RR   Y 4 OA 1   (192) ,1 1   1  sC 3   gmB 1   RY 4 ROA 1   1    1  V  gg AP  mA 1 mB 1  Vin Ds  Transimpedance mode (TIM): 1  1  V ggR  1  A  LP mA 1 mB 1 in 1  (193) Iin Ds  ,1  sC  1 g 1 R  1  A  V RR (194) parasitics. BP  Y 4 OA 1  Iin Ds  Voltage mode (VM): ,1 1 sC  11   1  gg 1  A RR VLP mA 1 mB 11  Y 1 O B1   (188) Vin Ds  ,1 1 (195)  sC 3   1  A 1  Rin ,1 1  1  V RR  sC 3   gmB 11  A 1  HP  Y 4 OA 1  VBP  RY 4 ROA 1 (189) Iin   Ds V  in Ds 11 , 11  sC 1   ,  sC 1   RY 1 RO B1   RY 1 RO B1   ,1 1 11 (190)  sC 3   Rin  1  A 1  ,  sC 3   1  A 1   RY 4 ROA 1  (196) VHP  RY 4 ROA 1   1   1  gg 1 mB 1   mA Ds Vin    VBR   ,1 1 I Ds   sC 1   in RR  Y 1 O B1  11    sC 1   ,11 , RR  sC 3    1  A 1   Y 1 O B1   RY 4 ROA 1  (191)   ,1 1 1   1  sC    gg  3   mA 1 mB 1  RR   Y 4 OA 1  Rin   (197) VBR   11 ,1  Vin Ds   sC 3   gmB 1   RY 4 ROA 1  11    V  gg AP  mA 1 mB 1  Iin Ds  D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Where, 11 1 s C  s C  2  1, R  ,2  A  g Z  R Y1 OB1  R OB2  1 mB 8 (207) 1 ,1 1 ,1 1 IBR  gmB 2 gmB 3  1  Ds  sC  sC     1  3  RRRR Ds * R  Y 1 O B1  Y 4 OA 1  Vin  in (198) 11 1 11 1 , ,,  A 1  sC 3   gmB 1  s C1   2  21  s C RR  Y 4 OA 1  R Y1 R OB1  R OB2   1  1   gg I 2 Z mA 1 mB 1 XA 6 1 (208) 1  ,   11  s C2  gmB 2  gmB 2 gmB 3 I R Current mode (CM): AP  OB2    * R V Ds in in Ig 1 g 1 LP mB 2 mB 3    (199) Where, Iin Ds  1 ,1  ,11 ,1  s C  g Ds  sC  sC  2 mB 2   1  2  I  R OB2  (200) RR R BP  Y 1 O B1  O B2   in Ds I  ,  1 1  A  g (209)  1 sC 2  2  mB 2 ,1 1 ,1 R  O B2  s C  s C  1  1  2   1  I RR R (201)  gg HP  Y1 OB1  OB2  mB 2 mB 3  2 Ds Iin  and A gZ 2 mA 9 11 1 The above eqns. (153-209) show the effect of parasitics s C  s C   1,  ,2  RR R in the proposed mixed-mode filter, mixed-mode first  Y1 OB1  OB2  (202) shadow filter, and mixed-mode second shadow filter. 1 IBR  gmB 2 gmB 3 However, the effect of parasitic capacitance can be ne-  1  in Ds I  glected by suitably choosing the value of , 2 CC, and 1 11 1 s C  s C   1,  ,2  RR R  Y1 OB1  OB2  (203) ,1   11  s C  g 1  gg   2  mB 2 mB 2 mB 3 I R AP  OB2  I  in Ds Transadmittance mode (TAM):  1  I gg 1 LP mB 2 mB 3  (204)  * R V Ds in in 1 1 ,   s C2  gmB 2 I R (205) BP  OB2   V Ds * R  in in 11 1 s C  s C   1,  ,2  I RR R (206) HP  Y1 OB1  OB2  Figure 11: Non-ideal equivalent circuit of Fig. 7 with V Ds * R in  in parasitics. D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 C3 . While parasitic resistances, except RX, can be easily neglected as it is high valued in the order of few MO. 5 Comparative analysis The non-shadow mixed-mode universal filter is com­pared to state of the art in section 5.1, followed by sec­tion 5.2, wherein mixed-mode shadow filters are com­pared. 5.1 Comparison with the existing SIMO mixed-mode non-shadow universal filters Table 3 compares the proposed filter with the existing SIMO mixed-mode biquad non-shadow universal fil­ters. All the topologies use more than one active build­ing block except [11] and the proposed one. Moreover, more passive components are used in [3, 7, 10, 11] than in the proposed one. Further, one or more passive com­ponents are floating in [3, 6, 7, 10, 11]. The filters [3, 7, 8] require matching components to realize the responses. Additional circuitry is required to obtain the simultane­ous responses for the VM in [4, 5, 8, 10, 12], TAM in [3, 5-8, 11], CM in [3-7, 11, 12], and TIM in [4, 5, 12]. Inde­ pendent tuning of . and Q is not possible in [5, 6, 9, oo 10], and electronic tuning is not possible in [3, 7, 8, 10, 11]. The topologies [3, 6, 9] consume less power than this work. However most of the filters are found to be partially cascadable including the proposed ones, only ref. [8] is fully cascadable. 5.2 Comparison with the existing different modes of shadow filters Table 4 compares the proposed mixed-mode shadow filters with the existing literature. It is noted that there is no report of any mixed-mode shadow filters realiza­tion using the same topology in literature except the proposed one. The topologies [13-20] realize only VM responses. Similarly, topologies [21-28] realize only CM responses, and [29] realizes only TIM and TAM respons­es. Whereas the proposed mixed-mode shadow filter realizes all the mixed-mode filters without alteration of the topology. The comparison Table 4 is self-explanato­ry for other parameters and features. 6 Simulated results and discussions The functionality of the proposed non-shadow and shadow mixed-mode filters is verified through the Ca­dence virtuoso spectre circuit simulator using TSMC 180 nm technology. The DC biasing levels of FD-CCCTA are taken as V= 1.2 V, V= -1.2V, V = V = 0, I= 25 DD SS bpbnbias1 µA, I = 20 µA, and I = I = 50 µA. Table 1 gives the bias2B1A1 aspect ratios of the transistors. The non-shadow filter of Fig. 3 is implemented with passive components chosen as C= C= C= 1 pF, and R= 1 .. Fig. 12 shows the lay­ 1 2 3 in out of the mixed-mode universal filter of Fig. 3, which occupies an area of 158.5 µm x 76.3 µm. For the VM and TIM, the pre-layout and post-layout gain responses of LP, HP, BP, and BR are shown in Fig. 13 (a), and the gain and the phase responses of AP are shown in Fig. 13 (b). Similarly, the responses for CM and TAM are shown in Fig. 14. The calculated pole frequency and the qual- (a) (b) D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Table 3: Comparative analysis with the existing SIMO mixed-mode universal filters. D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 Table 4: Comparative analysis with the existing different modes of shadow filters. ity factor are 46.22 MHz and 1, respectively, while the simulated pre-layout and post-layout frequencies are 46.45 MHz and 45.98 MHz, respectively. The simulation for the mixed-mode shadow filter of Fig. 7 is performed with the same parameters as used for the non-shadow filter along with 21 21 IB..IA..µA, R1 = R2 = 1.38 k.. The simulated gain responses of = = 40 lated f0s are obtained as 46.45 MHz, 23.19 MHz, and 11.75 MHz vis-à-vis the calculated values of 46.22 MHz, LP, HP, BP, and BR are shown in Fig. 13 (a), and the gain 23.11 MHz, and 11.55 MHz, respectively. The simulated and the phase responses of AP are shown in Fig. 13 (b) BWs are 46.41 MHz, 23.42 MHz, and 11.86 MHz vis-à-for the VM and TIM. Similarly, the responses for CM and vis the calculated BWs are 46.68 MHz, 23.34 MHz, and TAM are shown in Fig. 14. The simulated f0 and the Q 11.66 MHz, respectively. Fig. 15 (b) shows the tunability 259 are 46.41 MHz and 1.1 vis-a-vis the calculated values of 46.41 MHz and 0.95. The tunability of f0 along with BW for constant Q can be obtained by varying C1 = C3 = C. The simulated respons­es for the mixed-mode shadow filter of Fig. 6, using C = 1 pF, 2 PF, and 4 pF, are shown in Fig. 15 (a). The simu­ D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 of the quality factor along with the gain of the BP filter .. by the variation of gain A, i.e., varying 2 in line with 1gmB (46). The resulted quality factors are 0.8, 1.1, 1.3 and the .. gains are -1.2 dB, +1.4 dB, +3.7 dB for IB21 =40 µA, 200 µA, and 400 µA, respectively. The performance of the circuit is affected due to the fabrication process and mismatch deviation which has been analysed for the BP output response. Monte Carlo (MC) simulation for 200 runs is performed by consider­ing the deviation of standard parameters of MOSs. Fig. 16 (a) shows the MC results for the frequency response of BP in VM and TIM while Fig. 16 (b) shows for the same in CM and TAM. Fig. 17 (a) shows the histogram plot of the distribution of samples for center frequency in VM and TIM which results the standard deviation as 2.7 MHz. While, Fig. 17 (b) shows for the CM and TAM which results the standard deviation of 2.9 MHz. (a) (b) (a) (b) Figure 17: Statistical results of Monte Carlo simulation for BP output response (a) VM and TIM (b) CM and TAM. The PVT analysis has also been done for the Fast Fast (FF), nominal, and Slow Slow (SS) corners. Voltage has been varied in the range of 1.2 V ± 10 %. Whereas, tem­peratures have been taken as -40 oC, 27 oC, and 125 oC for the FF, nominal, and SS corners, respectively. Fig. 18 (a) shows for all the three corners which results in the centre frequencies of 53.5 MHz, 46.41 MHz, 40.32 MHz in the FF, nominal, and SS corners, respectively in the VM and TIM mode. While, Fig. 18 (b) shows the similar results in CM and TAM mode which gives the centre fre­quencies as 55.7 MHz, 46.41 MHz, 39.97 MHz. The measure of %THD (% total harmonic distortion) for the HP and LP mixed-mode shadow filter as a function of the input signal is shown in Fig. 19. The %THD varia­tion is less than 5% for VM and CM filters up to 600mA and 1000mA, as shown in Fig. 19 (a) and Fig. 19 (b), re­spectively. 7 Conclusions This paper presents a novel mixed-mode universal filter using a single active building block, FD-CCCTA, a new D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 variant of FD-CCII, and three capacitors. All the standard responses such as LP, BP, HP, BR, and AP are obtained in all VM, CM, TIM, and TAM modes. It is then extended to shadow filters to add flexibility in the orthogonal tuning of filter parameters, such as pole frequency (.), quality factor (Q), and the tuning of the filter’s gain. The first proposed shadow-filter circuit realizes the VM and CM UFs, while the second shadow filter realizes UFs for all the four modes, such as CM, TAM, VM, and TIM. To the best of the authors’ knowledge, there is no reporting of a mixed-mode shadow filter in literature. The theoreti­cal results are verified using TSMC 180 nm technology in Cadence Virtuoso Spectre. 8 Conflict of interest The authors declare that there is no conflict of interest for this paper. Also, there are no funding supports for this manuscript. 9 References 1. M. Parvizi, “Design of a new low power MISO mul­ti-mode universal biquad OTA-C filter,” Int. J. Elec, vol. 106, no. 3, pp. 440-454, 2018,. 2. M. Kumngern, P. Suwanjan, K. Dejhan, “Electroni­cally tunable voltage-mode universal fi lter with single-input five output using simple OTAs,” Int. J. Elec, vol. 100, no. 8, pp. 1118-1133, 2012. 3. B. Chaturvedi, J. Mohan, A. Kumar, “A new versa­tile universal biquad configuration for emerging, signal processing applications,” J.Circ. Syst. Comp, vol. 27, no. 12, pp. 1-28, 2018.. 4. H. P. Chen, W. S. Yang, “Electronically tunable cur­rent controlled current conveyor transconduct­ance amplifier-based mixed-mode biquadratic filter resistorless and grounded capacitors,” Appl. Sci, vol. 7, pp. 1-22, 2017. 5. M.A. Ibrahim, “Design and analysis of a mixed-mode universal filter using dual-output operational transconductance amplifiers (DO-OTAs). Proceed­ings of the international conference on computer and communication engineering, pp. 915-918, 2008. 6. M. Kumngern, S. Junnapiya, “Mixed-mode univer­sal filter using OTAs,” Proceedings of the 2012 IEEE D.Singh et al.; Informacije Midem, Vol. 52, No. 4(2022), 239 – 262 international conference on cyber technology in automation, control and intelligent systems, pp. 119-122, 2012. 7. W.B. Liao, J.C. Gu, “SIMO type universal mixed-mode biquadratic filter.” Indian Journal of Engi­neering & Materials Sciences, vol. 18, pp. 443-448. 8. S. Minaei, M.A. Ibrahim, “A mixed-mode KHN-biquad using DVCC and grounded passive ele­ments suitable for direct cascading,” Int. J. Circ. Theor. Appl, vol. 37, no. 7, pp. 793-810, 2009. 9. M. Parvizi, A. Taghizadeh, H. Mahmoodian, Z.D. Kozehkanani, “A low-power mixed-mode SIMO universal gm-C filter,” J. Circ. Syst. Comp, vol. 26, no. 10, pp. 1-16, 2017. 10. N.A. Shah, M.A. Malik, “Multifunction mixed-mode filter using FTFNs,” Analog Integr. Circ. Sig. Process, vol. 47, pp. 339-343, 2006. 11. C.N. Lee, C.M. Chang, “Single FDCCII-based mixed-mode biquad filter with outputs,” AEU-Int. J. Elec­tron. Comm., vol. 63, pp. 736-742, 2009. 12. L. Zhijun, “Mixed-mode universal fi lter using MC­CCII,” AEU-Int. J. Electron. Comm, vol. 63, pp. 1072­1075, 2009. 13. M.T. Abuelma’atti, N. Almutairi, “New CFOA-based shadow bandpass filter,” In: 15th international con­ference on electronics, information, and commu­nications, 2016. 14. M.T. Abuelma’atti, N. Almutairi, “New voltage-mode bandpass shadow filter,” In: 13th interna­tional multi-conference on systems, signals & de­vices, pp. 412-415, 2016. 15. M.T. Abuelma’atti, N.R. Almutairi, “New current-feedback operational amplifier based shadow fi l-ters,” Analog Integr. Circ. Sig. Process, vol. 86, pp. 471-480, 2016. 16. R. Anurag, R. Pandey, N. Pandey, M. Singh, M. Jain, “OTRA based shadow filters,” Annual IEEE India Conference, 2016. 17. P. Huaihongthong, A. Chaichana, P. Suwanjan, S. Siripongdee, W. Sunthonkanokpong, P. Supavarasu­wat, W. Jaikla, F. Khateb, “Single-input multiple-out­put voltage-mode shadow filter based on VDDDAs,” AEU-Int. J. Electron. Commun, vol. 103, pp. 13-23. 18. F. Khateb, W. Jaikla, T. Kulej, M. Kumngern, D. Kubanek, “Shadow filters based on DDCC,” IET Cir­cuits Devices Syst, vol. 11, pp. 631-637, 2017. 19. S.C. Roy,“Shadow filters: a new family of electronically tunable filters,” IETE J. Edu, vol. 51, pp. 75-78, 2010. 20. A. Yesil, F. Kacar, “Band-pass filter with high qual­ity factor based on current diff erencing transcon­ductance amplifier and current amplifi er” AEU-Int. J. Electron. Commun, vol. 75, pp. 63-69, 2017. 21. A. Yesil, F. Kacar, S. Minaei, “Electronically controllable bandpass filters with high quality factor and reduced capacitor value: an additional approach” AEU-Int. J. Electron. Commun. vol. 70, pp. 936-943, 2016. 22. M. Atasoyu, H. Kuntman, B. Metin, N. Herencsar, O.Cicekoglu, “Design of current-mode class 1 fre-quency-agile filter employing CDTAs,” European conference on circuit theory and design, 2015. 23. M. Atasoyu, B. Metin, H. Kuntman, N. Herencsar, “New current-mode class 1 frequency-agile fi lter for multi-protocol GPS application,” Elektronika IR Elektrotechnika, vol. 21, pp. 35-39, 2015. 24. D. Nand, N. Pandey, “New confi guration for OFCC-based CM SIMO filter and its application as shadow filter,” Arab. J. Sci. Eng. Vol. 43, pp. 3011-3022, 2018. 25. N. Pandey, R. Pandey, R. Choudhary, A. Sayal, M. Tripathi, “Realization of CDTA based frequency agile filters,” IEEE international conference on sig­nal processing computing and control, 2013. 26. N. Pandey, A. Sayal, R. Choudhary, R. Pandey, “De­sign of CDTA and VDTA based frequency agile fi l-ters,” Adv. Electron, pp. 1-14, 2014. 27. D. Singh, S.K. Paul, “Realization of current-mode universal shadow filter,” AEU-Int. J. Electron. Com-mun, vol. 117, pp. 153088, 2020. 28. D. Singh, S.K. Paul, “Improved current-mode bi-quadratic shadow universal filter,” Inf. MIDEM-J. Microelectron. Electron. Compon. Mater, vol. 52, no. 1, pp. 51-66, 2022. 29. D. Nand, N. Pandey, V. Bhanoo, A. Gangal, “Opera­tional floating current conveyor based TAM & TIM shadow filter,” Proceedings of 4th international conference on computer and management ICCM, pp. 103-115, 2018. 30. F. Gur, F. Anday, “Simulation of a novel current-mode universal filter using FDCCIIs,” Analog In-tegr. Circ. Sig. Process, vol. 60, pp. 231-236, 2009. 31. Y. Lakys, A. Fabre, “Shadow filters-new family of second order filters,” Electron Lett, vol. 46, no. 4, pp. 985-986, 2010. 32. V. Biolkova, D. Biolek, “Shadow filters for orthogo­nal modification of characteristics frequency and bandwidth,” Electronics Letters, vol. 46, no. 12, pp. 830-831, 2010. 33. M. Kumngern, T. Nonthaputha, F. Khateb, “Arbitrary waveform generators using current-controlled current conveyor transconductance amplifi er and current conveyor analog switches,” J. Circ. Syst. Comp, vol. 28, no. 11, pp. 1950179, 2019. Copyright © 2022 by the Authors. This is an open access article dis­tributed under the Creative Com­mons Attribution (CC BY) License (https://creativecom-mons.org/licenses/by/4.0/), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Arrived: 31. 07. 2022 Accepted: 24. 11. 2022 Original scientific paper https://doi.org/10.33180/InfMIDEM2022.405 Electronic Components and Materials Vol. 52, No. 4(2022), 263 – 269 User Offloading using Hybrid NOMA in Next-generation Heterogeneous Network Deepa Palani1, Merline Arulraj2 1Department of Electronics and Communication Engineering, Sethu Institute of Technology, Virudhunagar District, India 2Department of Electronics and Communication Engineering, Sethu Institute of Technology, Virudhunagar District, India Abstract: Millimeter wave (mmWave) enabled Heterogeneous network (Hetnet) has become ubiquitous because of the great demand of mobile network applications. Non–Orthogonal multiple access (NOMA) bids a desired possible assistance, for example condensed inactivity with great consistency, enhanced spectrum efficiency and considerable affinity. NOMA is envisioned to be used with small cells enabled with mmWave environment. This work proposes an ubiquitous connectivity between users at the cell edge and offloading macro cell so as to provide features the macro cell itself cannot cope with, such as extreme changes in the required user data rate and energy efficiency.The amount of inter-cell and performance is analyzed in the boundary and in the midpoint of the cell. It shows a reduction in outage possibility of 90% for cell center user (CCU) and 48% for cell edge user (CEU). Thereby alleviating dead zones and energy efficient support is shown for transmission using carrier sensing NOMA. Keywords: Heterogeneous network; millimeter wave; Non-Orthogonal multiple access; Small cells Uporabniška razbremenitev z uporabo hibridnega NOMA v heterogenem omrežju naslednje generacije Izvlecek: Heterogeno omrežje (Hetnet) z milimetrskim valovanjem (mmWave) je postalo vseprisotno zaradi velikega povpraševanja po mobilnih omrežnih aplikacijah. Neortogonalni veckratni dostop (NOMA) ponuja želeno možno pomoc, na primer zgošceno neaktivnost z veliko doslednostjo, izboljšano ucinkovitostjo spektra in veliko naklonjenostjo. NOMA naj bi se uporabljal z majhnimi celicami, ki jih omogoca okolje mmWave. To delo predlaga vseprisotno povezavo med uporabniki na robu celice in razbremenitev makrocelice, da se zagotovijo funkcije, ki jih makrocelica sama ne more obvladati, kot so ekstremne spremembe zahtevane hitrosti prenosa podatkov uporabnikov in energetske ucinkovitosti. Pokaže se zmanjšanje možnosti izpada za 90 % za uporabnika v središcu celice (CCU) in 48 % za uporabnika na robu celice (CEU). S tem se ublažijo mrtve cone in prikaže energetsko ucinkovita podpora za prenos z uporabo zaznavanja nosilcev NOMA Kljucne besede: Heterogeno omrežje; milimetrski valovi; neortogonalni veckratni dostop; majhne celice * Corresponding Author’s e-mail: venkatdeepa129@gmail.com 1 Introduction Fifth generation (5G) cellular networks require up to ways - orthogonal frequency division multiple access about a thousand times (1000x) more surface capacity (OMA) [2] and non-orthogonal multiple access (NOMA) than current long-term networks. In view of its [3]. By exploiting the multi-user diversity gain, system benefits sub-channel allocation is adapted using the throughput can be exploited in OMA [4]. It relies on the orthogonal frequency division multiple access (OFDM) known channel state information of all sub-channels multiplexing technique. [1] Achieving wireless multiple and accordingly sub-channel allocation is done for access on OFDM-based systems is implemented in two only one user. Regardless of the throughput system, How to cite: D.Palani et al., “User Offloading using Hybrid NOMA in Next-generation Heterogeneous Network", Inf. Midem-J. Microelectron. Electron. Compon. Mater., Vol. 52, No. 4(2022), pp. 263–269 D.Palani et al.; Informacije Midem, Vol. 52, No. 4(2022), 263 – 269 OMA has a disadvantage when communicating at very high data speeds. It does not allow frequency reuse in a cell [5], since a subcarrier in an OFDMA cell is allocated to only one user, which limits the sum data rate of the cell significantly. Unlike OFDMA, NOMA technique allocates a subcarrier to more than one user at a time in a cell, which ensures higher throughput due to sub-carrier reuse and therefore NOMA is considered as a major enabler for the next-generation heterogeneous networks. [6,7]The convergence of fixed and mobile access networks can be resolved by the NOMA access network. The system should allow the transmission of numerous types of broadband telecommunication traffic since the end user desires wireless broadband access. [8] The congestion is reduced in traditional wireless networks by offloading with small cells in Hetnet. 1.1 Motivation In order to improve the system performance, it is very stringent to use a single technology, e.g. Connectivity and the data-rate. NOMA utilizes the knowledge of superposition coding and successive interference cancellation in the transmitter and receiver. [9]. NOMA works in conjunction with various radio interface technologies, such as multi-cell scenarios, millimeter wave [10] and reconfigurable antenna systems [11], and other Internet of things platforms. NOMA technology is used with beamforming technology to improve the device to device communication in the small cell environment [12]. NOMA accomplishes this by multiplexing various transmitted signals into a stream of single signals by taking advantage of the power domain area. In view of the user centric point, successive interference cancellation (SIC) is the technique to receive the desired signal information and the remaining signals are disposed to remove interference. The request for execution of SIC is controlled by the expanding user channel state information (CSI) [13], i.e., the user is considered as the strong cell center user with high channel gain, this part is known as a cell center user (CCU), to unravel and offset data of the low-gain users, this part alluded to as a cell edge user (CEU). As of now, the utilization of NOMA in Hetnet includes device- to-device spectrum allocation and power control.Significantly three methodologies are utilized. In the cooperative NOMA near users (i.e CCU) is utilized to assist a distant user (i.e CEU). The second strategy is to introduce fairness among users by using design variables such as weighted sum-rate and the last strategy is to maintain the minimum requirement of the CEU. Therefore NOMA is treated as a transfer to help far users in wireless communication.[14] Therefore, the first challenge is to obtain the CSI of the co-channel users. There is strong residual interference among NOMA users without the knowledge of CSI of the interference, reliable communications is questionable. The second challenge is to remove the interference from the different signals of the co-channel users. A novel Interference cancellation technique is necessary to feat information of and suppresses the interference from all the overlapping symbols of the co-interference. It is accomplished using spectrum reuse techniques with NOMA known as user-pairing algorithm.The intra-cell interference is reduced and limited by using spectrum reuse techniques with Hybrid NOMA known as user-pairing algorithm. The contributions are as follows, - The user-pairing algorithm is applied to cancel out co-channel interference thereby achieving increased sum rate - Analyzing outage probability for the CCU and CEU thereby achieving offl oading macrocells. - A novel interference cancellation is exploited to reduce intracell and intercell interference thereby achieving spectral effi ciency. 2 System model A two-tier hetnet is considered where the macro and small base stations (BSs) are situated in the focal point of the cell and its inclusion region is a circle of range R. The BS serves Cell User equipment’s (CUEs) which are consistently circulated in the inclusion region. The precoding can be utilized to control the obstruction caused among the CUEs. The uplink system model for the nth user access is shown in Fig.1, where small cell environment is shown with cluster of users categorized into CCU and CEU. N is the total number of users in the cell, and N = 2M. The total number of antennas employed at the BS is represented as M. User equipment (UE) is attached with a single antenna. The different bandwidths allocated to UE in the OMA and hybrid NOMA schemes are demonstrated in Fig.2. The hybrid NOMA is served for different users irrelevant to position simultaneously in the same frequency. The femtocell user equipments (fCUE)are indicated in sub-channel which is assisted by femto BSs and indicated as macrocell user equipments (mCUE) in sub-channel which are assisted by macro BSs. The channel fading coefficients of UEs are assumed as ffff f h . h . h . h . ˆˆˆ . h (1) 1 234N ff Where h ˆˆ . h represents the channel fading 1N coefficient of strong signals in ascending order. Thus, in the given scenario, the received signals of the cluster group of uplink (UL) users, y, are given by, D.Palani et al.; Informacije Midem, Vol. 52, No. 4(2022), 263 – 269 N f ff y hx . px  n (2)  iN iN ll iN l . 1, li Where x  pfxf iN iNiN is the desired signal term transmitted symbol and ff power assigned to UE. l pxl is the interference from the other symbols niN denoted as the additive white Gaussian noise (AWGN) with variance s2.[15] 2.1 Path loss Model The mmWave propagation environment is modeled by widely adopted distance dependent path loss model and is given as Lr . 10  log r ° (3)   mm mm In the given equation . = 32.4 + 20log (fc) where fc represents the mmWave carrier frequency and the distance is represented as r. .mm is the shadow fading in mmWave link. The path loss exponent is denoted as a. [16] Figure 1: The System model of uplink NOMA-Hetnet Figure 2: Energy efficient NOMA based HetNets [15] NOMA uses a successive interference cancellation (SIC) technique to correctly demodulate signals at the receiver since it can allocate a subchannel to several users. The goal of differentiating users is achieved by SIC’s ability to remove interference in a certain order based on the power of various users. We presume that every terminal is fully aware of the channel state data (CSI). The uplink channel estimate in time division mode can provide with transmitter-side channel estimation, which can be used to obtain receiver-side channel estimation. Additionally, since one of our optimization goals for the given system is to maximise total harvested energy, we can infer that gathered energy somehow falls below the receiver saturation threshold. 2.2 Performance metrics The performance of mmWave enabled Hetnet is analyzed based on the metrics as Signal to Interference plus Noise Ratio (SINR), followed by Sum-Rate of an UE and Energy efficiency. Successive detection is carried out in descending order; The SINR is given as ff px 2 L  i N i N h i Nf mm r SINRiN  2 hf . N pf ˜ 2 (4) l N lN li 1  The achievable rate of NOMA UL user is denoted as ff px 2 L  i N i N h i Nf mm r  2 N (5) RiN  Blog 2  1  hf f 2 plN ˜  l N .li 1   where B is the bandwidth of mmWave enabled hetnets. The conventional OMA sum rate is obtained as 2 ff px Lmm r  i N i N h i Nf 2 N  (6) RiN  B / N log 2  1  f 2  p ˜  h l Nf .li 1 lN    The important parameter metric is energy effi ciency (EE) of the Hetnet. The objective is to target the decreased power consumption so as to increase the overall EE of a network.EE is maximized by optimizing the power allocation coeffi cients. Acheivabledatarate bps  EE  Joule (7) Total powerconsumption   s  Total Power consumption is derived as 1 f fff P . p .P (8) Total iN i N st p D.Palani et al.; Informacije Midem, Vol. 52, No. 4(2022), 263 – 269 The first component represented the dynamic power of femto BS which constituted the dissipation of power amplifi ers. The term #!" denoted the p 0,1 f power amplifi er effi ciency of the femto BS. is the iN fair power allocation ratio. The second component represents static power which is consumed by the transmitted signals and operating components. 3 Hybrid NOMA user-pairing algorithm Multiple users are supported by NOMA simultaneously in the same frequency in hybrid NOMA. Hybrid NOMA is a combined technique of OMA and NOMA,the users per carrier is increased indefinitely without compromising sum rate. A user-pairing primarily based totally suboptimal scheme. It is a proposed hybrid NOMA where different access schemes are combined with NOMA to facilitate the deployment. A user-pairing algorithm for Hybrid NOMA is served in different time slots to offl oad macrocell and to assure the quantity of accessed users. Hybrid NOMA user-pairing algorithm is derived to perform multiplexing of more than two users in the same frequency carrier to accommodate more users in hetnet. The user-pairing is done for two scenarios based on the distances of UEs. The user is considered as strong and weak according to the cell position. Near users is a strong one whereas far users are weaker. In order to offload the macro cell, small cells are assigned to provide ubiquitous connectivity using a user-pairing algorithm. This work aims to maximise the system’s overall energy efficiency while reducing the system’s energy consumption. As a result, this formula expresses the link between the system sum rate and overall power consumption. We also take the base station with the energy harvesting unit into account when allocating the subchannel and power resource. Hybrid NOMA User-Pairing Algorithm Initialize a) Set of users Ui . . 1,2,3 . . N . b) Set of sub-channels S . . 1,2,3 .. . K . c) Set the sub-channel power f allocation coeffi cient . iN d) Set of power values Ensure The number of split sub-channels k, the user-pairing strategy a) Near-Far user pairing (N-F) b) Near-near, far-far user pairing (N-N, F-F) Sort Sort transmission powers in ascending order. Assume P1< P2< ::: > > 1234 Choose Power allocation coeffi cient f 1 Ui . .. iN Obtain Sum Rate End if Here the far user is given the higher fraction of power whereas the near users are given lower fraction of power. But the cumulative result should not be ff exceeded 1. For example .. 0.75 and .. 0.25 i 1 i 2 for two users are assigned. The higher fraction of power is given to the far users. The distances are assumed for far user is d1=1000 meters and near user is d2=500 meters. 4 Results and discussion Simulation results are furnished to assess the overall performance of hybrid NOMA Hetnet.The overall performance of user-pairing hybrid NOMA schemes has been explored in which we have CCU and CEU ,i.e three UEs in a cell. The parameters are listed in table 1. Table 1: Evaluation Parameters Parameter Default Value Macro Transmit Power 46 dBm Femto Transmit Power 18 dBm Macrocell radius 500 m Femtocell radius 50 m Number of fCUE in each femtocell 3 Number of mCUE in macrocell 3 Shadowing standard deviation 10 dB System Bandwidth 100 MHz mmWave carrier frequency 28 GHz Noise power spectral density -174 dBm/Hz The transmit power of macrocells and femtocells and their coverage radius are listed. The total number of macro CUEs and femto CUEs are listed as well. However we have evaluated the scenario by increasing the number of UEs. The proposed user-pairing algorithm offl oading CEUs and fair power allocation solutions is evaluated through hybrid NOMA user-pairing algorithm and compared with the OMA strategies. Fig. 3, is portrayed D.Palani et al.; Informacije Midem, Vol. 52, No. 4(2022), 263 – 269 sum capacity in terms of their transmit Signal to noise Ratio (SNR). It is clear from Fig. 3 that hybrid NOMA scheme provides better sum-rate when compared to other schemes in perfect SIC mode. This performance is due to their distinct channel conditions, when the near strongest user (CCU) is paired with the far weak user (CEU), an achievable good sum rate is arrived. The improvement is not significant when pairing near-near and far-far, but still the sum rate performance is better compared to TDMA. The Single carrier NOMA (SC-NOMA) is not up to the level when compared to TDMA and hybrid NOMA. This is due to the overloading of users in the same carrier. Figure 4: Sum rate comparison between proposed hybrid NOMA scheme with fixed power allocation. Power allocation factor is the next important parameter that has the peculiar impact in the design performance of hybrid NOMA; hence evaluation is based on the distance between near user and BSs respectively. In Fig. 4 increasing transmitted power produces better sum rate capacity in the midst of lower SNRs. Thereby near user’s power to far users’ powers raises interference, for this reason fixed power allocation cannot be increased arbitrarily. An unconstraint maximization problem is used to find the optimal power-splitting ratio that returns the maximum instantaneous system throughput. Fig. 5 shows that outage is increased when the Signal to Interference Ratio (SIR)increases. When the distance increases, the outage probability is also increased for the multiple input multiple output (MIMO)-NOMA system. The greater the distance leads to greater interference.To examine the impact on outage performance, the transmit SNR of the two users is set to be equal and varied from 0 to 60 dB. The fi gure shows that, at a power ratio of 15 dB, both users’ outage performance achieves an interference-limited fl oor at a transmit SNR of roughly 15 dB, and that additional increases in transmit SNR have no effect on the users’ outage probability performance. D.Palani et al.; Informacije Midem, Vol. 52, No. 4(2022), 263 – 269 as the power ratio between the users rises, and many of the first user’s symbol estimates are highly likely to be accurate. The cluster comparison is given in Fig.7.Hybrid NOMA is a beneficial strategy for uplink transmission in future wireless communications since it can solve a signifi cant issue with OMA-based techniques, which is to not allow frequency reuse inside one cell. The impact of power allocation on NOMA communications led to the development of the unique hybrid NOMA technique which can take use of both the near-far eff ect and frequency reuse. Figure 7: Throughput Comparison between proposed hybrid NOMA schemes with fixed power allocation. Table 2: Data rate comparison between NOMA and OMA Distance 50m 100m 150m NOMA mCUE Data rate 2.2 Mbit/s 2 Mbit/s 1.8 Mbit/s f CUE Data rate 1.4 Mbit/s 1.2 Mbit/s 1 Mbit/s OMA mCUE Data rate 1.8 Mbit/s 1.5 Mbit/s 1.2 Mbit/s f CUE Data rate 1 Mbit/s 0.8 Mbit/s 0.4 Mbit/s Table 2 provides various user data rate experience, which are taken into account. As the average distance increases between users, it can be observed that the performance of OMA declines linearly. Due to lower average SNR, the second and third users achieve less bits per symbol. This is a result of significant user interference. The performance steadily improves as the power ratio rises, although it is still significantly inferior to OMA. Figure 8 shows the average EE comparison between OMA and NOMA schemes.The performance advantage over Hybrid NOMA grows as the power ratio rises. This is because Hybrid NOMA enables simultaneous access by allusers to a subcarrier. While the feasible bits/ symbol for the near and far users are lowered due to the reduced average received SNR, other users’ co-channel interference is also reduced.However, the poorest user’s energy efficiency decreases to roughly 1.8Mbits/ joule when OMA is used.We can see the superiority in attaining EE while using user-pairing scheme. Figure 8: Average EE between proposed hybrid NOMA scheme with OMA. 5 Conclusion Millimeter wave (mmWave) enabled Heterogeneous network (Hetnet) has offered ubiquitous connectivity with the aid of Hybrid NOMA user pairing algorithm because of the great demand. Hybrid NOMA is envisioned with superior performance with small cells enabled with mmWave environment. This proposed work outage performance is shown and offl oading macro cells such as extreme changes in the required user data rate and energy effi ciency.The congestion is reduced in traditional wireless networks by offl oading with small cells in Hetnet. The outage performance is analyzed for the alluded users particularly in the cell edge weak users. It shows a decrease in outage probability of 90% for near users i.e cell center user (CCU) and 48% for far usersi.e cell edge user (CEU). Thereby alleviating dead zones and energy effi cient support is shown for transmission using carrier sensing NOMA. A major enabler for handling the enormous number of Internet of devices that will be deployed in the factory of the future is widely recognized as being NOMA. 6 References 1. NTT DOCOMO, “5G Concept and Technologies”, http:5gworkshop.hhi.fraunhofer.de/wp-content/ D.Palani et al.; Informacije Midem, Vol. 52, No. 4(2022), 263 – 269 uploads/2020/12/Globecom-2020-WS-on-5G­New-Air-Interface-NTT-DOCOMO.pdf, December 2020. 2. B. Clerckx, Y. Mao, et al, (2021) “Is NOMA effi cient in multi-antenna networks? A critical look at next generation multiple access techniques,” IEEE Open Journal of the Communications Society, vol. 2, pp. 1310–1343, 2021. 3. L. Sala¨un, M. Coupechoux, and C. S. Chen, (2020) “Joint subcarrier and power allocation in NOMA: Optimal and approximate algorithms,” IEEE Trans­actions on Signal Processing, vol. 68, pp. 2215– 2230, 2020. 4. SepehrRezvani, Eduard A. Jorswieck, et al,(2021) Optimal SIC Ordering and Power Allocation in Downlink Multi-Cell NOMA Systems” IEEE Access, rXiv:2102.05015 [cs.IT] 5. O. Maraqa, A. S. Rajasekaran, S. Al-Ahmadi, H. Yanikomeroglu, and S. M. Sait, (2020) “A survey of rate-optimal power domain NOMA with enabling technologies of future wireless networks,” IEEE Communications Surveys & Tutorials, vol. 22, no. 4, pp. 2192–2235, 2020. 6. Tabassum, Hina&Hossain, Ekram&Hossain, Md. (2016). Modeling and Analysis of Uplink Non-Or­thogonal Multiple Access (NOMA) in Large-Scale Cellular Networks Using Poisson Cluster Pro­cesses. IEEE Transactions on Communications. PP. 10.1109/TCOMM.2017.2699180. 7. Zeng, Ming &Yadav, Animesh&Dobre, Octavia &Tsiropoulos, Georgios& Poor, H. Vincent. (2017). On the Sum Rate of MIMO-NOMA and MIMO­OMA Systems. IEEE Wireless Communications Let­ters. PP. 1-1. 10.1109/LWC.2017.2712149. 8. Batagelj Boštjan, Pavlovic, Leon, Naglic, Luka, Tomažic, Sašo. (2011) Convergence of fi xed and mobile networks by radio over fi bre technol­ogy.  Inf. MIDEM, jun. 2011, Vol. 41, no. 2, pp. 144­149. http://www.midem-drustvo.si/Journal%20 papers/MIDEM_41(2011)2p144.pdf. 9. S. Zhang et al., “A Dynamic Power Allocation Scheme in Power-Domain NOMA using Actor-Critic Reinforcement Learning,” 2018 IEEE/CIC International Conference on Communications in China (ICCC), 2018, pp. 719-723, https://doi.org/10.1109/ICCChina.2018.8641248. 10. N. C. Luong, D. T. Hoang, S. Gong, D. Niyato, P. Wang, Y. Liang, and D. I. Kim, “Applications of deep reinforcement learning in communications and networking: A survey,” IEEE Communications Surveys & Tutorials, vol. 21, no. 4, pp. 3133–3174, 4th Quarter 2019. 11. Y. Sun, J. Zhou, Q. Cao, and S. Li, “Precoder design in statistical CSI aided non-orthogonal multiple access,” IEEE Access, vol. 6, pp. 16 484–16 492, Feb. 2018. 12. Y. Liu, Z. Ding, M. Elkashlan, and J. Yuan, “Non-or­thogonal multiple access in large-scale underlay cognitive radio networks,” IEEE Trans. Veh. Tech-nol., vol. 65, no. 12, pp. 10 152–10 157, 2016 13. V.-D. Nguyen, H. D. Tuan, T. Q. Duong, O.-S. Shin, and H. V. Poor, “Joint fractional time allocation and beamforming for downlink multiuser MISO systems,” IEEE Commun. Lett., vol. 21, no. 12, pp. 2650–2653, 2017. 14. Q. Zhang, Q. Li, and J. Qin, “Robust beamform­ing for nonorthogonal multiple-access systems in MISO channels,” IEEE Transactions on Vehicular Technology, vol. 65, no. 12, pp. 10 231–10 236, Dec. 15. Zhang, Haijun et al. “Energy Effi cient Resource Management in SWIPT Enabled Heterogeneous Networks With NOMA.” IEEE Transactions on Wire­less Communications 19 (2020): 835-845. 16. Samuel Amalorpava Mary Rajee and ArulrajMer-line. Machine intelligence technique for blockage effects in next-generation heterogeneous net­works. Radioengineering, 29(3), 2020 Copyright © 2022 by the Authors. This is an open access article dis­tributed under the Creative Com­mons Attribution (CC BY) License (https://creativecom-mons.org/licenses/by/4.0/), which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Arrived: 27. 06.2 022 Accepted: 23. 11. 2022 Dobitnika prestižnih mednarodnih nagrad Electronic Components and Materials Vol. 52, No. 4(2022), 271 – 272 Dobitnika uglednih mednarodnih nagrad Winners of Prestigious International Awards Profesorica dr. Barbara Malic, predsednica Društva MIDEM, in profesor dr. Marko Topic, castni predsed­nik Društva MIDEM, sta v letu 2022 prejela ugledni mednarodni nagradi za izjemne prispevke in vrhunske dosežke na strokovnih podrocjih, ki jih pokriva naše društvo. V imenu Društva MIDEM jima iskreno cestitamo in želimo veliko novih raziskovalnih izzivov in uspehov! prof. dr. Janez Krc podpredsednik Društva MIDEM PROF. DR. BARBARA MALIC, DOBITNICA NAGRADE ZDRUŽENJA IEEE ZA PODROCJE FEROELEKTRIKOV Vodja Odseka za elektron­sko keramiko Instituta „Jožef Stefan“, prof. Barbara Malic, je dobitnica nagrade IEEE Ferroelectrics Recognition Award za leto 2022, ki jo podeljuje odbor za feroe­lektrike pri društvu Ultra­sonics, Ferroelectrics and Frequency Control Society (UFFC-S) v okviru zveze IEEE za dosežke na podrocju ra­ziskav feroelektrikov. Prof. Malic je nagrado prejela za izjemen prispevek k razjas­nitvi odnosov med kemijskimi in fizikalnimi lastnostmi feroelektricne keramike. V raziskavah feroelektricne ke­ramike na osnovi svincevih perovskitov in okolju prijaz­nejših alkalijskih niobatov se je posvecala razumevanju kemijskih procesov, ki potekajo med sintezo in sintran­jem keramike, in so osnova za nacrtovanje mikrostruk­ture in posledicno njihovih funkcijskih lastnosti. Nagra­da je veliko priznanje raziskovalnemu delu prof. Malic in potrditev odlicnosti raziskav feroelektrikov, ki Odsek za elektronsko keramiko že vrsto let uvršcajo med naj­pomembnejše raziskovalne skupine tega podrocja. Professor Dr. Barbara Malic, President of the MIDEM Soci­ety, and Professor Dr. Marko Topic, Honorary President of the MIDEM Society, received in 2022 prestigious interna­tional awards for outstanding contributions and achieve­ments in the fields of expertise covered by our Society. On behalf of the MIDEM Society, we congratulate them and wish them many new research challenges and suc­cesses! Prof. Dr. Janez Krc Vice-President of the MIDEM Society PROF. DR. BARBARA MALIC, WINNER OF THE IEEE PRIZE FOR FERROELECTRICS Prof. Barbara Malic, head of the Electron Ceramics De­partment at the Jožef Stefan Institute, is the recipient of the 2022 IEEE Ferroelectrics Recognition Award, present­ed by the Ferroelectrics Com­mittee of the Ultrasonics, Ferroelectrics and Frequency Control Society (UFFC-S) IEEE for achievements in the field of ferroelectrics re­search. Prof. Malic received the award for her outstand­ ing contribution to elucidat­ing the relationships between the chemical and physi­cal properties of ferroelectric ceramics. In her research on ferroelectric ceramics based on lead perovskites and the more environmentally friendly alkali niobates, she focused on understanding the chemical processes that occur during the synthesis and sintering of the ceramics and form the basis for the design of their microstructure and consequently their functional properties. The award is a great recognition of Prof. Malic’s research work and a confirmation of the excellence of ferroelectrics research, which has made the Electron Ceramics Department one of the most important research groups in this field for many years. Dobitnika prestižnih mednarodnih nagrad; Informacije Midem, Vol. 52, No. 4(2022), 271 – 272 PROF. DR. MARKO TOPIC, DOBITNIK EVROPSKE PROF. DR. MARKO TOPIC, WINNER OF BECQUERELOVE NAGRADE ZA PODROCJE THE EUROPEAN BECQUEREL PRIZE FOR FOTOVOLTAIKE PHOTOVOLTAICS Profesor dr. Marko Topic Professor Dr. Marko Topic s Fakultete za elektroteh- from the Faculty of Electri­niko Univerze v Ljubljani cal Engineering, University je letošnji dobitnik pres- of Ljubljana, is this year’s tižne evropske nagrade recipient of the prestig-“European Becquerel Prize ious “European Becquerel for Outstanding Merits in Prize for Outstanding Mer-Photovoltaics” za izjemne its in Photovoltaics” for his prispevke na podrocju exceptional contributions fotovoltaike. Ti vkljucuje- to the field of photovolta­jo njegovo akademsko in ics. These include his aca­znanstvenoraziskovalno demic and research work delo z razvojem številnih in developing a number opticnih in elektricnih si­mulatorjev ter merilnih sistemov, ki jih uporabljajo številni laboratoriji in industrija, njegovo izjemno anga­žiranost pri oblikovanju mocne fotovoltaicne skupnosti v Evropi in po svetu ter njegovo delo v vlogi predsedni­ka Evropske tehnološke in inovacijske platforme za fo­tovoltaiko (ETIP-PV.eu). Profesor Topic si ves cas svojega delovanja prizadeva za spodbujanje izmenjave znanja in ljudi, podpira prisotnost industrijskega PV-sektorja v EU ter zagotavlja pomoc številnim znanstvenikom in inženirjem, dejavnim tako v akademskih krogih kot v fotovoltaicni industriji. Uradna podelitev nagrade je bila na 8. svetovni konferenci o fotovoltaicni pretvorbi energije (WCPEC-8) 26. septembra 2022 v Milanu. of optical and electrical simulators and measurement systems used by many laboratories and industries, his great commitment to building a strong community of photovoltaic research­ers in Europe and worldwide, and his work as Chair of the European Technology and Innovation Platform for Photovoltaics (ETIP-PV.eu). The official award ceremony took place at the 8th World Conference on Photovoltaic Energy Conversion (WCPEC-8) on 26 th September 2022 in Milan. Call for papers Electronic Components and Materials Vol. 52, No. 4(2022), 273 – 273 MIDEM 2023 Chairs: Prof. Dr. Danjela Kušcer Assist. Prof. Dr. Kristina Žagar Soderžnik IMPORTANT DATES Abstract submission deadline: May 15, 2023 Acceptance notifi cation: June 26, 2023 Full paper submission deadline: July 31, 2023 Invited and accepted papers will be published in the Conference Proceedings. Detailed and updated information about the MIDEM Conferences, as well as for paper preparation can be found on http://www.midem-drustvo.si// 58th INTERNATIONAL CONFERENCE ON MICROELECTRONICS, DEVICES AND MATERIALS WITH THE WORKSHOP ON CHEMICAL SENSORS: MATERIALS AND APPLICATIONS September 27th – September 29th, 2023 Slovenia Announcement and Call for Papers GENERAL INFORMATION The 58th International Conference on Microelectronics, Devices and Materials with the Workshop on Chemical sensors: materials and applications continues a successful tradition of the annual international conferences organised by the MIDEM Society, the Society for Microelectronics, Electronic Components and Materials. The conference will be held from SEPTEMBER 27th – 29th, 2023. Topics of interest include but are not limited to: • Chemical sensors, • Novel monolithic and hybrid circuit processing techniques • New device and circuit design, • Process and device modelling, • Semiconductor physics, • Sensors and actuators, • Electromechanical devices, microsystems and nanosystems, • Nanoelectronics, • Optoelectronics, • Photovoltaic devices, • Electronic materials science and technology, • New electronic materials and applications, • Materials characterization techniques, • Reliability and failure analysis, • Education in microelectronics, devices and materials. ORGANIZER: MIDEM Society - Society for Microelectronics, Electronic Components and Materials, Slovenia CO-ORGANIZER: Republic of Slovenia, Ministry of Economic Development and Technology European Union, European Regional Development fund CONFERENCE SPONSORS: UL FE, UL FS, IJS, IMAPS, Slovenia Chapter; IEEE, Slovenia Section Boards of MIDEM Society | Organi društva MIDEM MIDEM Executive Board | Izvršilni odbor MIDEM President of the MIDEM Society | Predsednik društva MIDEM Prof. Dr. Barbara Malic, Jožef Stefan Institute, Ljubljana, Slovenia Vice-presidents | Podpredsednika Prof. Dr. Janez Krc, UL, Faculty of Electrical Engineering, Ljubljana, Slovenia Dr. Iztok Šorli, Mikroiks d.o.o., Ljubljana, Slovenia Secretary | Tajnik Olga Zakrajšek, UL, Faculty of Electrical Engineering, Ljubljana, Slovenia MIDEM Executive Board Members | Clani izvršilnega odbora MIDEM Prof. Dr. Slavko Bernik, Jožef Stefan Institute, Slovenia Assoc. Prof. Dr. Miha Cekada, Jožef Stefan Institute, Ljubljana, Slovenia Prof. DDr. Denis Ðonlagic, UM, Faculty of Electrical Engineering and Computer Science, Maribor, Slovenia Prof. Dr. Leszek J. Golonka, Technical University, Wroclaw, Poljska Prof. Dr. Vera Gradišnik, Tehnicki fakultet Sveucilišta u Rijeci, Rijeka, Croatia Mag. Leopold Knez, Iskra TELA, d.d., Ljubljana, Slovenia Mag. Mitja Koprivšek, ETI Elektroelementi, Izlake, Slovenia Asst. Prof. Dr. Gregor Primc, Jožef Stefan Institute, Ljubljana, Slovenia Prof. Dr. Janez Trontelj, UL, Faculty of Electrical Engineering, Ljubljana, Slovenia Asst. Prof. Dr. Hana Uršic Nemevšek, Jožef Stefan Institute, Ljubljana, Slovenia Dr. Danilo Vrtacnik, UL, Faculty of Electrical Engineering, Ljubljana, Slovenia Supervisory Board | Nadzorni odbor Prof. Dr. Franc Smole, UL, Faculty of Electrical Engineering, Ljubljana, Slovenia Prof. Dr. Drago Strle, UL, Faculty of Electrical Engineering, Ljubljana, Slovenia Igor Pompe, retired Court of honour | Castno razsodišce Darko Belavic, Jožef Stefan Institute, Ljubljana, Slovenia Dr. Miloš Komac, retired Dr. Hana Uršic Nemevšek, Jožef Stefan Institute, Ljubljana, Slovenia Informacije MIDEM Journal of Microelectronics, Electronic Components and Materials ISSN 0352-9045 Publisher / Založnik: MIDEM Society / Društvo MIDEM Society for Microelectronics, Electronic Components and Materials, Ljubljana, Slovenia Strokovno društvo za mikroelektroniko, elektronske sestavne dele in materiale, Ljubljana, Slovenija www.midem-drustvo.si