Original scientific paper
Informacije
^efMIDEM
A Innrnal of
Journal of Microelectronics, Electronic Components and Materials Vol. 45, No. 1 (2015), 47 - 56
Current-mode Quadrature Oscillator using CCCCTAs with Non-interactive Current Control for CO, FO and Amplitude
Adirek Jantakun
Department of Electronics and Telecommunication Engineering, Faculty of Engineering, Rajamangala University of Technology Isan, Khonkaen Campus, Khonkaen, Thailand
Abstract: This paper presents a current-mode quadrature oscillator with amplitude controllability. The proposed circuit consists of two current controlled current conveyor transconductance amplifiers, two grounded capacitors and one grounded resistor. The condition of oscillation, frequency of oscillation and amplitude of output current can be independently controlled with electronics method via DC bias current. In addition, the proposed oscillator has high impedance of outputs ports which facilitates easy driving an external load without additional current buffers. It also uses all grounded passive elements which suitable for implementation in the integrated circuit. The theoretical results are verified by PSPICE simulation and experimental results that well conform to the theoretical anticipation.
Keywords: Current-mode; Quadrature Oscillator; Current Controlled Current Conveyor Transconductance Amplifier; Grounded Elements
Kvadraturni oscilator v tokovnem načinu z uporabo CCCCTA z neinteraktivno kontrolo toka za CO, FO in amplitudo
Izvleček: Članek opisuje tokovno krmiljen kvadraturni oscillator z nadzorom amplitude. Predlagano vezje je sestavljeno iz dveh tokovno krmiljenih transkontuktančnih ojačevalnikov, dveh ozemljenih kondenzatorjev in ozemljenega upora. Frekvenca in amplituda oscilatorja je neodvisno nastavljiva preko vhodnega DC toka. Predlagan oscilator ima visoko izhodno impedanco, ki omogoča enostavno krmiljenje zunanjih bremen brez dodatnih virov. Zaradi ozemljenih pasivnih elementov je primerno za integrirana vezja. Teoretični izračuni so preverjeni v PSPICE in eksperimentalno potrjeni.
Ključne besede: tokovni način; kvadraturni oscillator; tokovno krmiljen transkonduktančni ojačevalnik; ozemljeni elementi
' Corresponding Author's e-mail: mradirek@hotmail.com
1 Introduction
An oscillator which provides two sinusoidal signals with 90° phase difference, or well known as quadrature oscillator, has been continuously proposed since it plays an important role and has been widely applied in various applications such as communication, instrumentation, measurement and signal processing, etc [14]. Especially in communication systems, the sinusoidal oscillator is frequently used to generate the carrier signal for modulation system [5-6] such as AM, FM, ASK etc. Normally, the oscillator would generate the signal without amplitude controllability. So in some applica-
tions which need the exact amplitude, the amplifier circuit is required, causing the complicated circuit. Recently, the portable electronics devices which run on compact batteries have been necessary to reduce voltage power supply and power consumption. These requirements call for the development of current-mode circuit designs due to their potential advantages such as inherently wide bandwidth, higher slew-rate, greater linearity, wider dynamic range, simple circuitry and low power consumption [7-8].
Several quadrature oscillators have been proposed in the literature [9-29]. A lot of them focus on the use of electronically adjustable active building blocks (ABB), such as, current controlled current conveyor (CCCII), current conveyor transconductance amplifier (CCTA), current controlled current conveyor transconductance amplifier (CCCCTA), differential voltage current conveyor transconductance amplifier (DVCCTA), current follower transconductance amplifier (CFTA) and current controlled current follower transconductance amplifier (CCCFTA). Some parameters (such as transconductance or parasitic resistance) of these ABBs are tuned by external current or voltage which is easy to control by microcontroller or microcomputer. The review of these oscillator topologies is as follows. The oscillators in Ref. [9-29] use grounded capacitor which is attractive for integrated circuit implementation. In Ref. [9-10], the electronic tuning of frequency of oscillation (FO) by adjusted parasitic resistance Rx is achieved. However, the condition of oscillation (CO) is controlled by current gain control, which is not so common [30-31]. CC-CIIs based sinusoidal oscillator was introduced in [11].
The magnitude of sinusoidal output current can be electronically adjusted as well as the frequency of oscillation, but this oscillator circuit employs three CCCIIs and the control of CO is done by adjusting the value of capacitor which is unconventional. The proposed oscillators in [12-14] exhibit advantages in electronic tuning of the FO by the transconductance gm, but the CO is controlled by change the value of passive resistor, which cause addition complication [30-31]. Although electronic adjusting can be used by digital potentiometer, but it is difficult for implementation [31]. The FO in [12, 19-24] is electronically adjusted by the transconductance gain gm. Some oscillators in [17, 25], the FO and CO cannot independently controlled. The circuits in [18-23, 26-27] can be electronically adjusted the CO and FO. The magnitude of sinusoidal output signal in [28] can be electronically adjusted but the FO and CO cannot be electronically adjusted. However, the amplitude of output signal in [9-10, 12-27, 29] cannot be electronically controlled. The comparison of proposed oscillator with previous work is shown in Table 1.
Table 1: The comparison of proposed oscillator with previous oscillators
Ref.	ABB	No. of ABB	No. of R+C	Electronic tune for both CO and FO	Independent tune of CO and FO	Output current with high impedance	Amplitude controllability
[9]	CG-CCCII	2	0+2	✓	✓	✓	x
[10]	CG-CCCII	2	0+2	✓	✓	x	x
[11]	CCCII	3	0+2	✓	x	✓	✓
[12]	CCTA		1+2	x	✓	x	x
[13]	CCTA		2+2	x	✓	✓	x
[14]	DVCCTA		2+2	x	✓	x	x
[15]	CCCCTA		0+2	x	✓	x	x
[16]	MO-CCCCTA		0+2	✓	x	✓	x
[17]	CG-CCTA		2+2	✓	x	x	x
[18]	CCTA	2	1+2	✓	x	✓	x
[19]	CCCCTA	2	0+2	✓	x	✓	x
[20]	CCCCTA	4	0+2	✓	✓	✓	x
[21]	MCDTA+ CFTA	2	0+2	✓	x	✓	x
[22]	CFTA	1	1+2	✓	x	✓	x
[23]	CCCFTA	2	0+2	✓	x	✓	x
[24]	ZC-CFTA	4	0+2	✓	✓	✓	x
[25]	CG-CFTA	2	0+2	✓	x	✓	x
[26]	CCCCT+ OTA	2	0+3	✓	x	✓	x
[27]	DO-CCCCTA	1	0+2	✓	x	✓	x
[28]	CDBA	2	3+2	x	x	✓	✓
[29]	DO-CCCCTA	1	2+2	✓	✓	✓	x
proposed circuit	CCCCTA	2	1+2	✓	✓	✓	✓
The purpose of this paper is to present the quadrature oscillator using CCCCTAs. The proposed circuit provides the following advantage features:
The current-output signal from high-impedance is easy to drive a load without using a buffering device [30-32]. The proposed circuit uses only grounded capacitors which are advantageous from the point of view of integrated circuit implementation [30-32].
The CO and FO can be electronically and independently controlled.
The amplitude of output current can be controlled via external bias current without effect of the CO and FO, which can provide the AM and ASK signal, widely used in communication systems [32].
Low active and passive sensitivities.
The workability of the proposed sinusoidal oscillator is confirmed via the PSPICE simulation and experiment.
2 Current Controlled Current Conveyor Transconductance Amplifier (CCCCTA)
Since the proposed circuit is based on CCCCTA [15], a brief review of CCCCTA is given in this section. It was modified from the first generation CCTA [8]. The characteristics of the ideal CCCCTA are represented by the following hybrid matrix:
ly		~ 0	0	0	0 "	
Vx		1	Rx	0	0	Ix
I., z.		0	1	0	0	Vo
Io		0	0	0	+S om _	V
(1)
For the CCCCTA implemented by a bipolar technology, the parasitic resistance R is given as
R =
2h,
and transconductance gm is given as
Ib2
Sm
2K
(2)
(3)
The circuit symbol and equivalent circuit of the CCCCTA are illustrated in Fig. 1(a) and (b), respectively
(a)
9Ib 2
O
>o
(b)

o
x
C»
z o
Figure 1: CCCCTA (a) schematic symbol (b) equivalent circuit
3 Proposed circuit configuration
The proposed quadrature oscillator is shown in Fig. 2. It consists of two CCCCTAs, two grounded capacitors and one grounded resistor. The quadrature output current ports are IO1, IO2 and IO3 is the output current with amplitude controllability. The use of grounded passive element is advantageous from the point of view of integrated circuit implementation [30-32]. Moreover, it is found that the oscillator provides high output impedance which can directly drive a load without buffering devices [30-32]. Using (1) and doing routine circuit analysis, the system characteristic equation can be expressed as
1 1
R1 Rx2
Ci C1C2RX1
= 0
(4)
From (4), it can be seen that the proposed circuit can produce oscillations if the condition of oscillation is fulfilled
R1 = Rx.
(5)
If the above condition of oscillation is satisfied, the circuit produces oscillation with frequency of
x
s
^osc =
Sml
CC2 Rxi
(6)
I02
ClSm1Rx
V
C2
(10)
Substituting the parasitic resistance Rx and transcon-ductance gm as respectively shown in (2) and (3) into (5) and (6), the CO becomes
I =
Substituting the parasitic resistance Rx and transcon-ductance gm as respectively shown in (2) and (3) into (10), the ratio between output current IO2 and IO1 becomes
2R,
and the FO is obtained as
fosc
2nV
^Bll^B 2 C1C2
(7)
(8)
Ir.
Ci Ib
V4C2 Ib
(11)
It is found that the CO can be electronically controlled through DC bias current without effect of the FO. Also, the FO can be electronically tuned through IB1 or IB2 without effect of the CO. Consequently, this circuit enables non-interactive current control for both the CO and the FO.
It is obvious seen from (11) that the tune of IB2 or IB1 for controlling the frequency of oscillation causes change of amplitude IO2 and IO1 during tuning process. When amplitude of IO2 or IO1 increases the voltage at notes Vy1 and Vz1 also increase too. This phenomenon will increase the THD if amplitude reaches high levels due to the limits of dynamical range of CCCCTA. However, this can be alleviated by simultaneously changing IB2 and Ib1 (IB2=IB1).
It is found that the phase difference of output current IO1 and IO2 is 90 However at oscillation state (M=Mosc), the relationship of IO2 and IO1 in (9) is changed to
zc
Figure 3: The non-ideal CCCCTA
^^^^ JQ22	^^

CC

r _r
Figure 4: The internal construction of CCCCTA

1
be electronically controlled by without effect of CO and FO. Therefore, if IB4 is a modulating signal, AM and ASK signal can be obtained at IO3. In addition, the output current IO3 port is high-impedance. It is easy to drive a load without using a buffering device.
The sensitivities of active and passive for frequency of oscillation are low as shown in (13)
^osc _ 1 C^osc
Sg.. 1 _ 2'	R„ 1
(13)
4 Non-ideal Analysis
Considering to voltage and current tracking errors, the CCCCTA properties can be written as
		~ 0	0	0	0 "	"v;
vx		Y	R.	0	0	h
Iz		0	a	0	0	v;
_ lo _		0	0	0		
(14)
Where	/(1 + s/a)Y), a_ao /(1 + s/a)a) and
ß_ßo /(1 + s/mß). They are frequency-dependence of non-ideal of voltage and current gains, respectively. 0}Y, aa and coß are dc non-ideal bandwidths with ideality equal to infinity. Jo, ao and ß^ are dc non-ideal gains with ideality equal to unity. Moreover, the effect of parasitic resistances and capacitances are also taken into account. The Cy, Cz and Co are parasitic capacitors at y, z and o terminals, respectively, with ideality equal to zero. The parasitic resistors R^, Rz, Ro at y, z and o terminals with ideality equal to infinity. These parasitic components of the CCCCTA are shown in Fig. 3.
Re-analysis the circuit in Fig. 2, the characteristic equation is modified to
s +s

CC*
_0 (15)
C2 C C ^
Then, the CO and FO of the proposed circuit become
(16)
Gz' + G* - G.L _ 0
C* C* C* And
"V
ßYraa1G1 gm1 + -
c;C;
(17)
Where C* _ Cj + C^j + Cy2 + + , C2 _ C2 + C^^
G_ - G _ -1-
G; _ Gy, + Gy2 + G-o, + Gz2 + Gr, , 1 R1 , x1 Rx1
0x2 ^^^	Gy1 ^^^ Gy2	G_1 _- 1
G _
R. 2, 1
R
z' Rz1 and
Gz 2 _
Ry 2 ,
R
R
5 Simulation Results
To prove the performances of the proposed oscillator, the PSPICE simulation was performed for examination. The BJT technology was simulated by using the parameters of the PR200N and NR200N bipolar transistors of ALA400 transistor array from AT&T. Fig. 4 depicts the schematic description of the CCCCTA used in the simulations with + 1.8 V power supplies.
20
(a)
20

o
-20
			'\l°2 P{\
			
92
94
96
Time(^s)
98
100
(b)
Figure 5: The waveforms of sinusoidal oscillator (a) transient state (b) steady state
The sinusoidal oscillator was designed with C1 = 1 nF, C2 = 0.47 nF, R1 = 1 k Q, IB1 = 58 ^A, IB2 = 115 ^A and IB4 = 10 ^A. From (7), the bias current IB3 was set nearby Vt/2R1 to start-up the oscillation, which is IB3 = 10 ^A. The power dissipation of this case is about 4.26 mW. This yields oscillation frequency of 630 kHz. Fig. 5 (a)
10^
1.0^ ■|^100n 10n
1.0n
1					
			fosc=630kHz THD(In1)=1.00% THD(In2)=2.21%		
A	^In,				
					
					
					
1.0	2.0	3.0	4.0
Frequency^Hz)
Figure 6: The frequency spectrum of signal in Fig. 5.
200
i 100
120
140
160
180
200
Time(^s)
Figure 7: The results of operation of AM
and (b) show simulated output waveforms in transient and steady state, respectively. Fig. 6 shows simulated output spectrum, where the total harmonic distortions (THD) of output current and are about 1.00 % and 2.21 %, respectively.
The simulated result of the proposed circuit serving as an AM generator at IO3 is shown in Fig. 7, where IB4 was triangular signal with a 50 kHz frequency. It is confirmed that the proposed circuit can easily generate AM signal.
The electronically adjustment of the FO can be demonstrated in Fig. 8 (a), while keeping IB2 = 115 ^A and tuning bias current IB1 from 10 ^A-200 ^A. Similarity, Fig. 8 (b) shows the simulation of FO, which remains IB1 = 58 ^A and adjusts IB2 from 20 ^A-190 ^A. It is clear seen that the simulation results are in accordance with the theoretical analysis as shown in equation (8). The error of the oscillation frequency stems from the nonideal parameters as depicted in the Section IV. The amplitude of output current IO1 can be exhibited in Fig. 9, when tuning bias current IB1 and IB2.
□ Theoretical O Simulation
N
10 20 40 60 80 100 120 140 160 180 200
1b1(mA)
(a)
□ Theoretical O Simulation
N 900
pH
20 40 60 80 100 120 140 160 180 190
(b)
Figure 8: The frequency of oscillation against bias current (a) Ib1 (b) I^^
circuit for experiment is illustrated in Fig. 10. It was constructed by using commercial ICs, which are AD844A [33] and LM13700N [34]. The values of components in experimental circuit were chosen as C^ = 1 nF, C2 = 0.47 nF, R^ = 1.1 kO, R^j = 1 kO (R^j was used to measure the output current by an oscilloscope). The intrinsic resistances R.^^ and R2 are realized by OTA-based grounded resistance simulator with following value
2F
R =
L
From (16) the CO and FO can be rewritten as
2Vt
1b3 =
R1
(16)
(17)
6 Experimental Results
To prove the theoretical analysis of proposed oscillator, the experimental results are shown in this section. The
And
f =
J osc
1
1B11B 2
CjC2
107
10
10'
10-
10"
10 20 40 60 80 100 120 140 160 180 200
Ibi(^A)
(a)
10-4r-
10"
10"
20 40 60 80 100 120 140 160 180 190 Ib2(uA)
(b)
Figure 9: The amplitude of against bias current (a)
Ib, (b) IB2
The relationship of Vo2 and Vo1 is as follows:
I V
V _ ^BV O1
2VtSC2
(19)
Similarly, the amplitude of output voltage VO3 is shown in (20)
J R V
2V
(20)
The circuit in Fig. 10 was biased with = 60 ^A, = 115 ^A, IB3 = 56 ^A, IB4 = 60 ^A and ±5 V power supplies. An oscilloscope Rigol model DS1046B [35] was used to measure the output waveforms. The waveform of output signals are shown in Fig. 11 (a) where the FO is about 321 kHz. The calculated value of FO from (18) is about 370.84 kHz. The deviation of FO is suffered from parasitic elements of the CCII, OTA and parasitic element of oscilloscope. Moreover, the parasitic influences of the CCII and OTA are shown in Table 2 and 3, respectively. Fig. 11 (b) exhibits the signals of VO1 and VO2 , which are the phase difference of 90° Furthermore, the frequency spectrum of output signals by use signal analyzer Agilent Technologies model N9000A [36] shown in Fig. 12, that is the FO is about 320.6 kHz. The parameters in Fig. 12 can be calculated the THD of VO1, VO2 and VO3 are about 1.73 %, 1.53 % and 1.31 %, respectively. These THD (VO1, VO2 and VO3) are close to the simulation, which are about 1.50 %, 1. 27 % and 2.17 %, respectively.
To generate the AM signal, IB4 is feed as triangular waveform with 25 kHz of frequency. The voltage of current converter in Fig. 13 is used for this case. Fig. 14 shows the AM signal. It is clearly seen that this result confirms the theoretical analysis as shown in (20).
Figure 10: The oscillator circuit for experimental test
Table 2: The parasitic influences of CCII (AD844A) [33]
Port	Resistance	Capacitance
y	7 MQ	2 pF
x	6 5Q	2 pF
z	3 MQ	4.5 pF
Table 3: The parasitic influences of OTA (LM13700N) [34]
Port	Resistance	Capacitance
Input	100 kQ(IB=100 ^A)	5.2 pF
output	11 MQ(Ib=100 ^A)	4.6 pF
(a)
(b)
Figure 11: (a) The waveform of output signals (VO1, VO2 and VO3), (b) The waveform between VO1 and VO2
7 Conclusion
Current-mode quadrature oscillator using CCCCTAs and grounded elements has been presented. The proposed circuit consists of two CCCCTAs, two grounded capaci-
Figure 12: The spectrum of output signal
Figure 13: The V to I converter circuit
STOP 10.00US m^
I o SS0mU
Freq=25:0kHž
	4	.....	....:.[[ ,: .p...:....:
:Upp=: 2.04U
CHI- 1.00U
50.0mU
Figure 14: The waveform of AM signal
tors and one grounded resistor. The condition and frequency of oscillation are independently adjusted by input bias current of CCCCTAs. Due to high-output impedances, it enables easy driving load without external current buffer. In addition, the amplitude of sinusoidal output signal can be electronically tuned. Moreover, it can provide the AM/ASK signals that are widely used in communication systems [32]. The PSpice simulation and experimental results well conform to the theoretical anticipation.
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Arrived: 18. 06. 2014 Accepted: 03. 12. 2014